Zero division duplexing mimo radio with adaptable rf and/or baseband cancellation

ABSTRACT

A intelligent backhaul radio is disclosed, which can operate by zero division duplexing for use in PTP or PMP topologies, providing for significant spectrum usage benefits among other benefits. Specific system architectures and structures to enable active cancellation of multiple transmit signals at multiple receivers within a MIMO radio are disclosed. Further disclosed aspects include the adaptive optimization of cancellation parameters or coefficients.

PRIORITY

The present application is a continuation application of U.S. patentapplication Ser. No. 14/572,725 filed Dec. 16, 2014, which is acontinuation application of U.S. patent application Ser. No. 14/108,200,filed on Dec. 16, 2013, (now U.S. Pat. No. 8,948,235 issued Feb. 3,2015) which is a continuation application of U.S. patent applicationSer. No. 13/767,796, filed on Feb. 14, 2013, (now U.S. Pat. No.8,638,839 issued Jan. 28, 2014) which is a continuation application ofU.S. patent application Ser. No. 13/609,156, filed on Sep. 10, 2012 (nowU.S. Pat. No. 8,422,540 issued Apr. 16, 2013) which claims priority toProvisional Application Ser. No. 61/662,809, filed on Jun. 21, 2012, andProvisional Application Ser. No. 61/663,461, filed on Jun. 22, 2012, theentireties of which are hereby incorporated by reference.

BACKGROUND

1. Field

The present disclosure relates generally to data networking and inparticular to a backhaul radio for connecting remote edge accessnetworks to core networks with advantageous spectrum usage.

2. Related Art

Data networking traffic has grown at approximately 100% per year forover 20 years and continues to grow at this pace. Only transport overoptical fiber has shown the ability to keep pace with thisever-increasing data networking demand for core data networks. Whiledeployment of optical fiber to an edge of the core data network would beadvantageous from a network performance perspective, it is oftenimpractical to connect all high bandwidth data networking points withoptical fiber at all times. Instead, connections to remote edge accessnetworks from core networks are often achieved with wireless radio,wireless infrared, and/or copper wireline technologies.

Radio, especially in the form of cellular or wireless local area network(WLAN) technologies, is particularly advantageous for supportingmobility of data networking devices. However, cellular base stations orWLAN access points inevitably become very high data bandwidth demandpoints that require continuous connectivity to an optical fiber corenetwork.

When data aggregation points, such as cellular base station sites, WLANaccess points, or other local area network (LAN) gateways, cannot bedirectly connected to a core optical fiber network, then an alternativeconnection, using, for example, wireless radio or copper wirelinetechnologies, must be used. Such connections are commonly referred to as“backhaul.”

Many cellular base stations deployed to date have used copper wirelinebackhaul technologies such as T1, E1, DSL, etc. when optical fiber isnot available at a given site. However, the recent generations of HSPA+and LTE cellular base stations have backhaul requirements of 100 Mb/s ormore, especially when multiple sectors and/or multiple mobile networkoperators per cell site are considered. WLAN access points commonly havesimilar data backhaul requirements. These backhaul requirements cannotbe practically satisfied at ranges of 300 m or more by existing copperwireline technologies. Even if LAN technologies such as Ethernet overmultiple dedicated twisted pair wiring or hybrid fiber/coax technologiessuch as cable modems are considered, it is impractical to backhaul atsuch data rates at these ranges (or at least without adding intermediaterepeater equipment). Moreover, to the extent that such special wiring(i.e., CAT 5/6 or coax) is not presently available at a remote edgeaccess network location; a new high capacity optical fiber isadvantageously installed instead of a new copper connection.

Rather than incur the large initial expense and time delay associatedwith bringing optical fiber to every new location, it has been common tobackhaul cell sites, WLAN hotspots, or LAN gateways from offices,campuses, etc. using microwave radios. An exemplary backhaul connectionusing the microwave radios 132 is shown in FIG. 1. Traditionally, suchmicrowave radios 132 for backhaul have been mounted on high towers 112(or high rooftops of multi-story buildings) as shown in FIG. 1, suchthat each microwave radio 132 has an unobstructed line of sight (LOS)136 to the other. These microwave radios 132 can have data rates of 100Mb/s or higher at unobstructed LOS ranges of 300 m or longer withlatencies of 5 ms or less (to minimize overall network latency).

Traditional microwave backhaul radios 132 operate in a Point to Point(PTP) configuration using a single “high gain” (typically >30 dBi oreven >40 dBi) antenna at each end of the link 136, such as, for example,antennas constructed using a parabolic dish. Such high gain antennasmitigate the effects of unwanted multipath self-interference or unwantedco-channel interference from other radio systems such that high datarates, long range and low latency can be achieved. These high gainantennas however have narrow radiation patterns.

Furthermore, high gain antennas in traditional microwave backhaul radios132 require very precise, and usually manual, physical alignment oftheir narrow radiation patterns in order to achieve such highperformance results. Such alignment is almost impossible to maintainover extended periods of time unless the two radios have a clearunobstructed line of sight (LOS) between them over the entire range ofseparation. Furthermore, such precise alignment makes it impractical forany one such microwave backhaul radio to communicate effectively withmultiple other radios simultaneously (i.e., a “point to multipoint”(PMP) configuration).

In particular, “street level” deployment of cellular base stations, WLANaccess points or LAN gateways (e.g., deployment at street lamps, trafficlights, sides or rooftops of single or low-multiple story buildings)suffers from problems because there are significant obstructions for LOSin urban environments (e.g., tall buildings, or any environments wheretall trees or uneven topography are present).

FIG. 1 illustrates edge access using conventional unobstructed LOS PTPmicrowave radios 132. The scenario depicted in FIG. 1 is common for many2^(nd) Generation (2G) and 3^(rd) Generation (3G) cellular networkdeployments using “macrocells”. In FIG. 1, a Cellular Base TransceiverStation (BTS) 104 is shown housed within a small building 108 adjacentto a large tower 112. The cellular antennas 116 that communicate withvarious cellular subscriber devices 120 are mounted on the towers 112.The PTP microwave radios 132 are mounted on the towers 112 and areconnected to the BTSs 104 via an nT1 interface. As shown in FIG. 1 byline 136, the radios 132 require unobstructed LOS.

The BTS on the right 104 a has either an nT1 copper interface or anoptical fiber interface 124 to connect the BTS 104 a to the Base StationController (BSC) 128. The BSC 128 either is part of or communicates withthe core network of the cellular network operator. The BTS on the left104 b is identical to the BTS on the right 104 a in FIG. 1 except thatthe BTS on the left 104 b has no local wireline nT1 (or optical fiberequivalent) so the nT1 interface is instead connected to a conventionalPTP microwave radio 132 with unobstructed LOS to the tower on the right112 a. The nT1 interfaces for both BTSs 104 a, 104 b can then bebackhauled to the BSC 128 as shown in FIG. 1.

The conventional PTP radio on a whole is completely unsuitable forobstructed LOS or PMP operation. To overcome, these and otherdeficiencies of the prior art, one or more of the inventors hasdisclosed multiple exemplary embodiments of an “Intelligent BackhaulRadio” (or “IBR”) in U.S. patent application Ser. No. 13/212,036, nowU.S. Pat. No. 8,238,318, and Ser. No. 13/536,927, which share a commonassignee to this invention and are hereby incorporated by reference intheir entirety into this invention.

Both conventional PTP radios and IBRs can be operated using timedivision duplexing (or “TDD”). FIG. 2 depicts a generic digital radio,not necessarily a conventional PTP radio or an IBR, in a TDDconfiguration. With TDD, only one of the radios amongst the two peersdepicted in FIG. 2 transmits at any given point in time according to atime schedule known to both radios. This enables both TDD radios totransmit and receive at the same frequency or within the same finitefrequency band. Thus, relative to FIG. 2, each radio transmits andreceives at a single nominal carrier frequency f₁, but when one radio inany given link (of which only one link is depicted in FIG. 2) istransmitting, then the other radio is receiving wherein neither radiotransmits and receives at the same point in time.

Both conventional PTP radios and IBRs can be operated using frequencydivision duplexing (or “FDD”). FIG. 3 depicts a generic digital radio,not necessarily a conventional PTP radio or an IBR, in a conventionalFDD configuration. With conventional FDD, each of the radios amongst thetwo peers depicted in FIG. 3 transmits at separate frequencies, f₁ andf₂, known to both radios but chosen such that frequency-selectivefilters within the transmitters and the receivers of each radio cansufficiently reject unwanted transmit noise and signal leakage withineach receiver. In FDD, both radios amongst the two peers depicted inFIG. 3 can transmit and receive simultaneously.

There are well-known advantages and disadvantages for each of TDD or FDDoperation for both the generic radios of FIGS. 2 and 3, or for IBRs orconventional PTP radios. Many advantages of TDD are related todisadvantages of FDD, or vice versa.

For example, in TDD antenna resources are easily shared between transmitand receive, which is a distinct advantage for TDD. Conversely, in FDDantenna resources are not easily shared and either separate antennas fortransmit and receive or a frequency duplexer is required to supportconventional FDD radio operation. For example, in utilizing TDD, theisolation of the receiver from the intentional or unintentional signalsof the transmitter is straightforward since transmission occurs onlywhen reception does not occur, and vice versa. Conversely, in utilizingFDD, the transmitter signals (both intentional and unintentional) mustbe highly isolated from the receiver channel, which is a disadvantagefor FDD such that conventional FDD radios have required multipleadditional filters relative to TDD as well as separate antennas orfrequency duplexers. For example in TDD, spectrum access is veryflexible as any portion of the radio spectrum that supports the desiredchannel bandwidth can theoretically be used and the duty cycle betweentransmit and receive can be varied easily, which is another distinctadvantage for TDD. Conversely, in FDD spectrum access channels fortransmit and receive need to be paired at frequency separationscompatible with the isolation requirements for conventional FDD radiooperation.

However, even conventional FDD radios have significant advantages overTDD despite the disadvantages described above. For example in FDD, withseparate antennas between transmit and receive, the losses betweeneither the power amplifier and the transmit antenna or the receiveantenna and the low noise amplifier are lower than that of a TDD radiowith a transmit/receive switch, which is an advantage for such an FDDradio. As another example in FDD, very high PHY and MAC efficiency canbe obtained in a backhaul radio with simultaneously achieving very lowlatency in transmission and reception without any significant bufferingat the network interface, which is a distinct advantage for FDD.Conversely, in TDD a trade-off is required between PHY and MACefficiency on one hand and minimizing latency and buffering on the otherhand. For example, in FDD all of the “non-antenna” resources, such asthe entire transmit path and the entire receive path are utilized at alltimes. Conversely, in TDD much or all of the transmit path may be idlewhen receiving and vice versa for the receive path when transmitting,which significantly underutilizes the resources within a TDD radio. Forexample, in FDD a low latency feedback channel to each transmitter isavailable because a peer FDD radio does not need to wait until suchtransmitter stops transmitting to send feedback information that mayenhance the link performance. Conversely, in TDD no feedback can occuruntil the transmitter stops and the radio goes to receive mode and thusfeedback latency must also be traded off against overall efficiency.

Recently, it has been proposed that certain types of radio systems thatoperate within a single channel or band may transmit and receivesimultaneously without all of the isolation circuitry and frequencyseparation of conventional FDD systems by adding a cancellationcapability as shown conceptually in FIG. 4. In such an exemplaryconfiguration, which is referred to herein as “Zero Division Duplexing”(or “ZDD”), each radio in a particular link transmits and receives atsingle nominal carrier frequency f₁ as is allowable for TDD, but cantransmit and receive simultaneously as is allowable for FDD Such a ZDDradio has many of the advantages of FDD, such as for example only, lowlatency and high efficiency, as well as many of the advantages of TDD,such as for example only, flexible spectrum access. Furthermore, a ZDDradio in theory can have at least twice the single-link spectralefficiency of either an FDD or TDD radio by utilizing the same frequencychannel simultaneously in both directions of a link. In someconfigurations, ZDD operation may be at two channels about a nominalfrequency f₁ such that no conventional isolation filtering at RF ispossible in which case the above referenced advantages still applyexcept for the spectral efficiency improvement. As shown in FIG. 4,exemplary ZDD radios have attempted to cancel unwanted transmitterleakage primarily by inverting the phase of an attenuated copy of thetransmit signal and summing it at the receiver to substantially cancelthe transmit signal leakage at the receiver antenna.

However, known techniques for implementing ZDD radios are completelyinadequate for an IBR, or many other high performance antenna arraybased radio systems, to operate in any ZDD mode whether “co-channel” or“co-band”.

SUMMARY

The following summary of the invention is included in order to provide abasic understanding of some aspects and features of the invention. Thissummary is not an extensive overview of the invention and as such it isnot intended to particularly identify key or critical elements of theinvention or to delineate the scope of the invention. Its sole purposeis to present some concepts of the invention in a simplified form as aprelude to the more detailed description that is presented below.

Some embodiments of the claimed invention are directed to backhaulradios utilizing zero division duplexing (ZDD) that are compact, lightand low power for street level mounting, operate at 100 Mb/s or higherat ranges of 300 m or longer in obstructed LOS conditions with lowlatencies of 5 ms or less, can support PTP and PMP topologies, use radiospectrum resources efficiently and do not require precise physicalantenna alignment. Radios with such exemplary capabilities as describedby multiple various embodiments are referred to herein by the term“Intelligent Backhaul Radio” (IBR).

According to aspects of the invention, an intelligent backhaul radio isdisclosed that includes a plurality of transmit RF chains to convertfrom a plurality of transmit chain input signals to a plurality ofrespective RF transmit chain signals; a plurality of adaptable RFtransversal filter sets to convert a plurality of signals respectivelyderived from the plurality of RF transmit chain signals to a pluralityof RF transmit leakage cancellation signals, wherein each adaptable RFtransversal filter set is comprised of a plurality of adaptable RFtransversal filters, each for filtering a respective one of saidplurality of signals respectively derived from the plurality of RFtransmit chain signals to provide a respective adaptable RF transversalfiltered signal, and a combiner for combining the plurality of adaptableRF transversal filtered signals within the filter set to produce one ofsaid RF transmit leakage cancellation signals; a plurality of RFcancellation combiners for respectively combining one of said pluralityof RF transmit leakage cancellation signals with one of said pluralityof RF receive signals to provide a plurality of RF receive chain inputsignals, and a plurality of receive RF chains to convert from saidplurality of RF receive chain input signals to a plurality of respectivereceive chain output signals; a cancellation controller, for adaptingeach of said plurality of adaptable RF transversal filters of one ormore of said plurality of adaptable RF transversal filter sets, whereinthe cancellation controller utilizes a plurality of RF transmit leakagemetrics respectively derived from each of said receive RF chainsassociated with each of said one or more adaptable RF transversalfilters sets being adapted by said cancellation controller; one or moredemodulator cores, wherein each demodulator core demodulates one or morereceive symbol streams to produce a respective receive data interfacestream; a frequency selective receive path channel multiplexer,interposed between the one or more demodulator cores and the pluralityof receive RF chains, to produce the one or more receive symbol streamsprovided to the one or more demodulator cores from the plurality ofreceive chain output signals; an antenna array that includes a pluralityof directive gain antenna elements; and one or more selectable RFconnections that selectively couple certain of the plurality ofdirective gain antenna elements to certain of the plurality of receiveRF chains, wherein the number of directive gain antenna elements thatcan be selectively coupled to receive RF chains exceeds the number ofreceive RF chains that can accept receive RF signals from the one ormore selectable RF connections; and a radio resource controller, whereinthe radio resource controller sets or causes to be set the specificselective couplings between the certain of the plurality of directivegain antenna elements and the certain of the plurality of receive RFchains.

According to other aspects of the invention an intelligent backhaulradio is disclosed that includes a plurality of transmit RF chains toconvert from a plurality of transmit chain input signals to a pluralityof respective RF transmit chain signals; a plurality of adaptable RFtransversal filter sets to convert a plurality of signals respectivelyderived from the plurality of RF transmit chain signals to a pluralityof RF transmit leakage cancellation signals, wherein each adaptable RFtransversal filter set is comprised of a plurality of adaptable RFtransversal filters, each for filtering a respective one of saidplurality of signals respectively derived from the plurality of RFtransmit chain signals to provide a respective adaptable RF transversalfiltered signal, and a combiner for combining the plurality of adaptableRF transversal filtered signals within the filter set to produce one ofsaid RF transmit leakage cancellation signals; a plurality of RFcancellation combiners for respectively combining one of said pluralityof RF transmit leakage cancellation signals with one of said pluralityof RF receive signals to provide a plurality of RF receive chain inputsignals, and a plurality of receive RF chains to convert from saidplurality of RF receive chain input signals to a plurality of respectivereceive chain output signals; a plurality of receive basebandcancellation combiners for combining a plurality of baseband transmitleakage cancellation signals with a respective one of said plurality ofreceive chain output signals to provide a plurality of basebandcancelled receive signals; a plurality of adaptable baseband transversalfilter sets to receive a plurality of signals respectively derived fromthe plurality of transmit chain input signals and provide a plurality ofbaseband transmit leakage cancellation signals respectively to each ofthe plurality of receive baseband cancellation combiners, wherein eachadaptable baseband transversal filter set is comprised of a plurality ofadaptable baseband transversal filters, each for filtering a respectiveone of said plurality of signals respectively derived from the pluralityof transmit chain input signals to provide a respective adaptablebaseband transversal filtered signal, and a combiner for combining theplurality of the adaptable baseband transversal filtered signals of thefilter set s to provide one of said baseband transmit leakagecancellation signals to one of said respective receive basebandcancellation combiners, a cancellation controller, for adapting each ofsaid plurality of adaptable RF transversal filters of one or more ofsaid plurality of adaptable RF transversal filter sets, wherein thecancellation controller utilizes a plurality of RF transmit leakagemetrics respectively derived from each of said receive RF chainsassociated with each of said one or more adaptable RF transversalfilters sets being adapted by said cancellation controller; and whereinsaid cancellation controller is additionally for adapting each of saidplurality of adaptable baseband transversal filters of one or more ofsaid plurality of adaptable baseband transversal filter sets, whereinthe cancellation controller utilizes a plurality baseband cancellationadaptation input signals derived directly or indirectly from each ofsaid receive RF chains or from said baseband combiners associated witheach of said one or more adaptable baseband transversal filter setsbeing adapted by said cancellation controller; one or more demodulatorcores, wherein each demodulator core demodulates one or more receivesymbol streams to produce a respective receive data interface stream; afrequency selective receive path channel multiplexer, interposed betweenthe one or more demodulator cores and the plurality of receive basebandcancellation combiners, to produce the one or more receive symbolstreams provided to the one or more demodulator cores from the pluralityof baseband cancelled receive signals; an antenna array that includes aplurality of directive gain antenna elements; and one or more selectableRF connections that selectively couple certain of the plurality ofdirective gain antenna elements to certain of the plurality of receiveRF chains, wherein the number of directive gain antenna elements thatcan be selectively coupled to receive RF chains exceeds the number ofreceive RF chains that can accept receive RF signals from the one ormore selectable RF connections; and a radio resource controller, whereinthe radio resource controller sets or causes to be set the specificselective couplings between the certain of the plurality of directivegain antenna elements and the certain of the plurality of receive RFchains.

In some embodiments, the transmit leakage metric is an RSSI measurement.

In some embodiments, the transmit leakage metric is further derived fromsaid receive chain output signal, wherein the derivation of said RFtransmit leakage metric comprises a correlation with one or more signalsrelated to one or more of said transit chain input signals.

In some embodiments, the adapting by said cancellation controller ofeach of the plurality of adaptable RF transversal filters of one or moreof said plurality of adaptable RF transversal filter sets utilizes aniterative adaptation algorithm so as to minimize or otherwise optimizesaid wherein said RF transmit leakage metrics.

In some embodiments, the baseband cancellation adaptation input signalsare respectively derived said baseband cancelled receive signals.

In some embodiments, the baseband cancellation adaptation input signalsare respectively derived utilizing a correlation with one or moresignals related to one or more of said transit chain input signals, andwherein said adapting by said cancellation controller of each of theplurality of adaptable baseband transversal filters of one or more ofsaid plurality of adaptable baseband transversal filter sets utilizes aniterative adaptation algorithm so as to minimize any remaining transmitleakage signal from the baseband cancelled receive signals.

In some embodiments, the baseband cancellation adaptation input signalscomprise signals related said transit chain input signals, and furthercomprise one or more of said receive chain output signals, and whereinsaid adapting by said cancellation controller of each of the pluralityof adaptable baseband transversal filters of one or more of saidplurality of adaptable baseband transversal filter sets utilizes aclosed form calculation utilizing said baseband cancellation adaptationinput signals.

In some embodiments, the closed form calculation involves a leastsquares or MMSE calculations.

In some embodiments, the combiner of one or more of said adaptable RFtransversal filter sets is integral to one or more of said RFcancellation combiners.

The In some embodiments, the combiner of one or more of said adaptablebaseband transversal filter sets is integral to one or more of saidreceive baseband cancellation combiners.

According to an aspect of the invention, an intelligent backhaul radiois disclosed that includes a plurality of transmit RF chains to convertfrom a plurality of transmit chain input signals to a plurality ofrespective RF transmit chain signals; a plurality of transmit RFreference receive chains respectively coupled, directly or indirectly,to the output of said plurality of transmit RF chains, to convert aplurality of signals respectively derived from said respective RFtransmit chain signals to respective baseband sampled RF transmitreference signals; a plurality of receive baseband cancellationcombiners for combining a plurality of baseband sampled RF transmitleakage cancellation signals with a respective one of a plurality ofreceive chain output signals to provide a plurality of basebandcancelled receive signals; a plurality of first adaptable basebandtransversal filter sets to receive the plurality of baseband sampled RFtransmit reference signals and provide a plurality of baseband sampledRF transmit leakage cancellation signals respectively to each of theplurality of respective receive baseband cancellation combiners, whereineach first adaptable baseband transversal filter set is comprised of aplurality of first adaptable baseband transversal filters, each forfiltering a respective one of said plurality of baseband sampled RFtransmit reference signals to provide a respective first basebandfiltered signal, and a combiner for combining the plurality ofrespective first baseband filtered signals within the filter set toprovide one of said plurality of baseband sampled RF transmit leakagecancellation signals to one of said respective receive basebandcancellation combiners; a plurality of RF cancellation combiners forrespectively combining one of a plurality of up-converted basebandtransmit leakage cancellation signals with one of said plurality of RFreceive signals to provide a plurality of RF receive chain inputsignals, and a plurality of receive RF chains to convert from aplurality of said RF receive chain input signals to a plurality of saidrespective receive chain output signals; a plurality of cancellationup-converter chains, each to receive a respective one of said pluralityof baseband transmit leakage cancellation signals and respectivelyprovide one of said plurality of the up-converted baseband transmitleakage cancellation signals to a respective one of said plurality of RFcancellation combiners; a plurality of second adaptable basebandtransversal filter sets to receive a plurality of signals respectivelyderived from the plurality of transmit chain input signals and providethe plurality of baseband transmit leakage cancellation signalsrespectively to each of the plurality of respective cancellationup-converter chains, wherein each second adaptable baseband transversalfilter set is comprised of a plurality of second adaptable basebandtransversal filters, each for filtering a respective one of saidplurality of signals respectively derived from the plurality of transmitchain input signals to provide a respective second baseband filteredsignal, and a combiner for combining the plurality of the secondbaseband filtered signals of the filter set to provide one of saidplurality of baseband transmit leakage cancellation signals to one ofsaid respective cancellation up-converter chains, a cancellationcontroller, for adapting each of said plurality of second adaptablebaseband transversal filters of one or more of said plurality of secondadaptable baseband transversal filter sets, wherein the adaptationcontroller utilizes a second transmit leakage metric derived, directlyor indirectly, from each of said receive RF chains, or said receivebaseband cancellation combiners associated with each of said one or moresecond adaptable transversal filter sets being adapted by saidadaptation controller, wherein said cancellation controller isadditionally for adapting each of said plurality of said first adaptablebaseband transversal filters of one or more of said plurality of firstadaptable baseband transversal filter sets, wherein the adaptationcontroller utilizes a plurality of baseband cancellation adaptationinput signals derived directly or indirectly from each of said receiveRF chains, or receive baseband cancellation combiners associated witheach of said one or more first adaptable baseband transversal filtersets being adapted by said adaptation controller.

In some embodiments, the second RF transmit leakage metric is an RSSImeasurement.

In some embodiments, the second RF transmit leakage metric is furtherderived from said receive chain output signal, wherein the derivation ofsaid first or said second RF transmit leakage metric comprises acorrelation with one or more signals related to one or more of saidtransit chain input signals.

In some embodiments, the adapting by said cancellation controller ofeach of the plurality of first or second adaptable RF transversalfilters of one or more of said plurality of first or second adaptable RFtransversal filter sets utilizes an iterative adaptation algorithm so asto minimize or otherwise optimize said RF transmit leakage metrics.

In some embodiments, the baseband cancellation adaptation input signalsare respectively derived said baseband cancelled receive signals.

In some embodiments, the baseband cancellation adaptation input signalsare respectively derived utilizing a correlation with one or moresignals related to one or more of said transit chain input signals, andwherein said adapting by said cancellation controller of each of theplurality of second adaptable baseband transversal filters of one ormore of said plurality of second adaptable baseband transversal filtersets utilizes an iterative adaptation algorithm so as to minimize anyremaining transmit leakage signal from the baseband cancelled receivesignals.

In some embodiments, the baseband cancellation adaptation input signalscomprise signals related said transit chain input signals, and furthercomprise one or more of said receive chain output signals, and whereinsaid adapting by said cancellation controller of each of the pluralityof second adaptable baseband transversal filters of one or more of saidsecond plurality of adaptable baseband transversal filter sets utilizesa closed form calculation utilizing said baseband cancellationadaptation input signals.

In some embodiments, the closed form calculation involves a leastsquares or MMSE calculations.

In some embodiments, the combiner of one or more of said secondadaptable baseband transversal filter sets is integral to one or more ofsaid RF cancellation combiners

In some embodiments, the combiner of one or more of said first adaptablebaseband transversal filter sets is integral to one or more of saidbaseband cancellation combiners.

In some embodiments, the coupling of the transmit RF reference receivechains, directly or indirectly, to the output of said plurality oftransmit RF chains includes the coupling to and from one or more of thefollowing: a power amplifier, one more frequency selective RFcomponents, an RF switch fabric, an RF Front-end, a Front-endTransmission Unit, a low pass filter, a band pass filter, a notchfilter, a high pass filter, an equalizing filter, a duplexing filter,one or more radio frequency switch or switches, an RF coupler, an RFdivider, a Wilkinson divider or combiner, a splitter, a summer, acombiner, a BALUN, an RF circulator, an RF isolator, a transmissionline, a micro-strip line, an RF front end module, an antenna, adirective gain element, an antenna including the coupling of receivedsignals from other antennas, an antenna including signals reflected froma transmit antenna as a result of imperfect impedance matching, acomponent including a “Enable” input that causes substantially allactive circuitry to power down, a component including a “Enable” inputthat causes a substantial reduction in RF energy.

In some embodiments, multiple nested or successive RF cancellationprocesses are utilized to increase the cancellation prior to anyintermediate frequency, analog or baseband cancellation processes.

In some embodiments, multiple nested or successive Intermediatefrequency cancellation processes are utilized to increase thecancellation prior to any analog or baseband cancellation processes.

In some embodiments, multiple nested or successive analog basebandcancellation processes are utilized to increase the cancellation priorto any baseband cancellation processes.

In some embodiments, multiple nested or successive digital basebandcancellation processes are utilized to increase the cancellation.

In some embodiments, crest factor reduction techniques are utilized withthe transmitter to reduce non-linear distortion.

In some embodiments, a frequency selective transmit equalizer isutilized so as to increase an isolation aspect between the transmitantenna array and the receive antenna array, or associated elements orcoupling ports; selected or collectively.

In some embodiments, the optimization of a frequency selective transmitequalizer includes metrics associated with a target receivingintelligent backhaul radio and the isolation aspects of the receiveantenna array of the current intelligent backhaul radio relative to thetransmit antenna array associated with the transmit equalizer.

In some embodiments, the frequency selective transmit equalizer utilizestransmit beam forming.

In some embodiments, adaptive transmitter beam forming is utilized so asto reduce requirements of cancellation, in on aspect including a reducedtime delay of the received RF coupling paths required to be canceled byan RF cancellation process, or the amplitude of signals related totransmitter to receiver coupling within a given delay spread.

In some embodiments, more antennas than receive chains are present andthe selection of the receive antennas is based upon a combination of ametric associated with the desired receive signals and a metricassociated with the isolation between the selected receive antennaelements from the transmit antenna array associated with the sameintelligent backhaul radio.

In some embodiments, more antennas than receive chains are present andthe selection of the receive antennas is based upon a combination of ametric associated with the capacity of the received signal from aseparate transmitting intelligent backhaul radio, and the impact to theresulting capacity at the current intelligent backhaul radio as a resultof interference from transmitted signals from the same intelligentbackhaul radio.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the potential for the receivechains receivers to be saturated by transmitter leakage at one or moreof the low noise amplifier, a RF selection switch, and an analog todigital coverer maximum input level.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal at RF.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal at an intermediatefrequency.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal at analog baseband.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal at digital baseband.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal, including an un-cancelabletransmitter noise.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter noise associated with transmitter signal leakage from thereceived signal.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal, and the ability to utilizea frequency selective transmit equalizer to satisfy a transmit capacityto a target intelligent backhaul radio.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal, and the ability to utilizea transmit automatic gain control to satisfy a transmit capacity to atarget intelligent backhaul radio.

In some embodiments, the impact to the capacity at the currentintelligent backhaul radio is based upon the ability to cancel thetransmitter leakage from the received signal, and the ability to utilizea transmit automatic gain control and a frequency selective transmitequalizer to satisfy a transmit capacity to a target intelligentbackhaul radio.

In some embodiments, a transmitter associated with the intelligentbackhaul controller having an automatic gain control wherein theadjustment of the automatic gain control is based at least in part uponthe remaining transmitter leakage signal level following a transmitterleakage cancellation process in one or more of the receivers.

In some embodiments, a transmitter associated with the intelligentbackhaul controller having an automatic gain control wherein theadjustment of the automatic gain control is based at least in part uponthe remaining transmitter leakage noise level following a transmitterleakage cancellation process in one or more of the receivers.

In some embodiments, a transmitter associated with the intelligentbackhaul controller having an automatic gain control wherein theadjustment of the automatic gain control is based at least in part uponthe remaining transmitter leakage signal level or noise level followinga transmitter leakage cancellation process in one or more of thereceivers taking into account a pre-determined level causing non-lineardistortion effects in one or more receiver components.

In some embodiments, the non-linear distortion is an analog to digitalconverter maximum receive level, or dynamic range.

In some embodiments, the non-linear distortion is maximum receive level,or dynamic range associated with a desired receiver sensitivity.

In some embodiments, weights for use with an RF cancellation are storedduring an initial factory calibration.

In some embodiments, weights for use with an RF cancellation are storedduring an initial factory calibration, wherein more antennas thenreceive chains are present and the weights are stored according to anindex associated with specific antenna selections.

In some embodiments, weights for use with an RF cancellation are storedfollowing optimization of the performance of the cancellation process.

In some embodiments, weights for use with an RF cancellation are storedfollowing optimization of the performance of the cancellation process,wherein more antennas then receive chains are present and the weightsare stored according to an index associated with specific antennaselections.

In some embodiments, more antennas then receive chains are present, andweights are stored according to an index associated with specificantenna selections.

In some embodiments, weights for use with an RF cancellation are storedfollowing the decision to change the selection of an antenna where anindex associated with the current antenna selection is utilized in theweight storage.

In some embodiments, weights for use with an RF cancellation areretrieved following the decision to change the selection of an antenna,where an index associated with the specific antenna selection isutilized in the weight retrieval.

In some embodiments, more antennas then receive chains are present, andthe weights are stored and retrieved according to an index associatedwith specific antenna selections.

In some embodiments, storing and retrieval of weights associated withone or more cancellation processes is performed based upon an event.

In some embodiments, storing and retrieval of weights for one or morecancellation processes is performed based upon a reselection ofantennas.

In some embodiments, only digital baseband cancellation of transmitterleakage signal is performed.

In some embodiments, the adaptation of the weights associated with asubsequent cancellation step is performed while the weights associatedwith a preceding transmitter cancellation step are held constant.

In some embodiments, the adaptation or calculation of the weightsassociated with a transmitter cancellation step is performed during aspecific time period coordinated by the radio resource controller.

In some embodiments, the radio resource controller of the currentintelligent backhaul radio coordinates one or more properties of thespecific time period with a respective intelligent backhaul radio havingsignal received by the current intelligent backhaul radio.

In some embodiments, the one or more properties of the specific timeperiod include the time or duration of the specific time period.

In some embodiments, the one or more properties of the specific timeperiod include the transmitted or received signal power levels ofsignals received by the current intelligent backhaul radio during thespecific time period.

-   -   a. In some embodiments, the one or more properties of the        specific time period include the transmitted wave forms received        by the current intelligent backhaul radio during the specific        time period.

In some embodiments, the intelligent backhaul radio utilizes common upconverting local oscillator signals.

In some embodiments, the intelligent backhaul radio utilizes common downconverting local oscillator signals.

In some embodiments, the intelligent backhaul radio utilizes common downconverting local oscillator signals and analog to digital samplingtiming signals.

In some embodiments, the intelligent backhaul radio includes receiversperforming zero division duplexing cancellation of the transmittersignals and performing spatial multiplexing among the set of receivers,and utilizes common down converting local oscillator signals.

In some embodiments, the intelligent backhaul radio includes receiversperforming zero division duplexing cancellation of the transmittersignals and performing spatial multiplexing among the set of receivers,and performs an RF cancellation and a digital baseband cancellationprocess.

In some embodiments, the RF cancellation processing includes twocancellation steps comprising a first RF cancellation followed by asecond RF cancellation.

In some embodiments, both RF cancellations are based upon signalsderived form a sample RF transmitter signals.

In some embodiments, one RF cancellation is based upon signals derivedfrom sampled RF transmitter signals and the other RF cancellation isbased upon up converted digital cancellation signals derived fromdigital baseband transmitter signals.

In some embodiments, the digital baseband cancellation process utilizescancellation signals sampled from the intelligent backhaul radiotransmitters at RF.

In some embodiments, the digital baseband process comprises a pluralityof cancellation steps a first cancellation step utilizing cancellationsignals sampled from the intelligent backhaul radio transmitters at RFand a second cancellation process utilizing cancellation signals derivedfrom digital baseband transmitter signals.

In some embodiments, the digital baseband process utilizes cancellationsignals sampled from the intelligent backhaul radio transmitters atdigital baseband.

In some embodiments, at least one transmitter non-linearity is simulatedat digital baseband utilizing digital baseband transmitter inputsignals, and utilized in a digital baseband cancellation process.

In some embodiments, the simulation of a transmitter non-linearity atdigital baseband utilizes a received signal to estimate parameters ofthe non-linearity.

In some embodiments, the simulation of a transmitter non-linearity atdigital baseband utilizes a metric following the cancellation process toestimate parameters of the non-linearity.

In some embodiments, the cancellation of a simulated transmitternon-linearity at digital baseband involves the removal of the digitalbaseband transmitter signal from the output of the non-linearityestimation process and the application of a digital filter estimating achannel response between the transmitter to receiver, prior to the stepof performing a cancellation of the estimated non-linearity distortionfrom a receiver signal.

In some embodiments, a plurality of the receiver chain input signals,receive antenna signals, RF transmit reference signals, or signals inputto a RF canceller combiner are combined in a predetermined gain, delay,or phase relationship so as to be separable in digital baseband.

In some embodiments, a plurality of the receiver chain input signals,receive antenna signals, RF transmit reference signals, or signals inputto a RF canceller combiner are combined in a predetermined gain, delay,or phase relationship so as to be separable in digital baseband where inthe signals derived from the RF transmitter reference signals areseparated and utilized in a cancellation process.

In some embodiments, each individual cancellation process includes one,multiple, or all of the components or steps associated with foregoingcancellation steps, processes or blocks.

Aspects of the current invention also include combinations andpermutations of the foregoing embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated into and constitute apart of this specification, illustrate one or more examples ofembodiments and, together with the description of example embodiments,serve to explain the principles and implementations of the embodiments.

FIG. 1 is an illustration of conventional point to point (PTP) radiosdeployed for cellular base station backhaul with unobstructed line ofsight (LOS).

FIG. 2 is an illustration of a generic TDD radio.

FIG. 3 is an illustration of a generic FDD radio.

FIG. 4 is an illustration of a generic ZDD radio.

FIG. 5 is an illustration of intelligent backhaul radios (IBRs) deployedfor cellular base station backhaul with obstructed LOS according to oneembodiment of the invention.

FIG. 6 is a block diagram of an IBR according to one embodiment of theinvention.

FIG. 7 is a block diagram of an IBR according to one embodiment of theinvention.

FIG. 8 is a block diagram of an IBR antenna array according to oneembodiment of the invention.

FIG. 9A is a block diagram of a Front-end Transmission Unit according toone embodiment of the invention.

FIG. 9B is a block diagram of a Front-end Reception Unit according toone embodiment of the invention.

FIG. 10 is a diagram of an exemplary horizontally arranged intelligentbackhaul radio antenna array.

FIG. 11 is a diagram of an exemplary vertically arranged intelligentbackhaul radio antenna array.

FIG. 12 is a block diagram of a portion of a ZDD enabled IBR accordingto one embodiment of the invention.

FIG. 13 is a block diagram of a Loop 1 ZDD canceller according to oneembodiment of the invention.

FIG. 14 is a block diagram of a Loop 2 ZDD canceller according to oneembodiment of the invention.

FIG. 15 is a block diagram of a Loop 3 ZDD canceller according to oneembodiment of the invention.

FIG. 16 is a block diagram of a portion of a ZDD enabled IBR including aZDD Canceller Loop coefficients generator according to one embodiment ofthe invention.

FIG. 17 is a block diagram of a portion of a ZDD enabled IBR including aLoop 1 (C1) and Loop 3 (C3D) cancellers according to one embodiment ofthe invention.

FIG. 18 is a block diagram of a portion of a ZDD enabled IBR including aLoop 2 (C2D) and Loop 3 (C3R) cancellers according to one embodiment ofthe invention.

FIG. 19 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR transmitter according to one embodiment ofthe invention.

FIG. 20 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR receiver according to one embodiment of theinvention.

FIG. 21 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR receive chain input according to oneembodiment of the invention.

FIG. 22 is a diagram of a further mathematical representation ofdepicted signals at a ZDD enabled IBR receive chain input according toone embodiment of the invention.

FIG. 23 is a diagram of a further detailed mathematical representationof depicted signals at a ZDD enabled IBR receive chain input accordingto one embodiment of the invention.

FIG. 24 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR receive chain digital output according toone embodiment of the invention.

DETAILED DESCRIPTION

FIG. 5 illustrates deployment of intelligent backhaul radios (IBRs) inaccordance with an embodiment of the invention. As shown in FIG. 5, theIBRs 500 are deployable at street level with obstructions such as trees504, hills 508, buildings 512, etc. between them. The IBRs 500 are alsodeployable in configurations that include point to multipoint (PMP), asshown in FIG. 5, as well as point to point (PTP). In other words, eachIBR 500 may communicate with more than one other IBR 500.

For 3G and especially for 4^(th) Generation (4G), cellular networkinfrastructure is more commonly deployed using “microcells” or“picocells.” In this cellular network infrastructure, compact basestations (eNodeBs) 516 are situated outdoors at street level. When sucheNodeBs 516 are unable to connect locally to optical fiber or a copperwireline of sufficient data bandwidth, then a wireless connection to afiber “point of presence” (POP) requires obstructed LOS capabilities, asdescribed herein.

For example, as shown in FIG. 5, the IBRs 500 include an Aggregation EndIBR (AE-IBR) and Remote End IBRs (RE-IBRs). The eNodeB 516 of the AE-IBRis typically connected locally to the core network via a fiber POP 520.The RE-IBRs and their associated eNodeBs 516 are typically not connectedto the core network via a wireline connection; instead, the RE-IBRs arewirelessly connected to the core network via the AE-IBR. As shown inFIG. 5, the wireless connection between the IBRs include obstructions(i.e., there may be an obstructed LOS connection between the RE-IBRs andthe AE-IBR).

FIGS. 6 and 7 illustrate exemplary embodiments of the IBRs 500 shown inFIG. 5. In FIGS. 6 and 7, the IBRs 500 include interfaces 604, interfacebridge 608, MAC 612, modem 624, channel MUX 628, RF 632, which includesTx1 . . . TxM 636 and Rx1 . . . RxN 640, antenna array 648 (includesmultiple antennas 652), a Radio Link Controller (RLC) 656, a RadioResource Controller (RRC) 660 and a ZDD Canceller 670. The IBR mayoptionally include an IBMS agent 700 as shown in FIG. 7. It will beappreciated that the components and elements of the IBRs may vary fromthat illustrated in FIGS. 6 and 7. Multiple exemplary embodiments of thecomponents and elements of the IBRs of FIGS. 6 and 7, except for ZDDCanceller 670, are disclosed in U.S. patent application Ser. No.13/212,036, now U.S. Pat. No. 8,238,318, and Ser. No. 13/536,927 andincorporated herein.

As described in greater detail in U.S. patent application Ser. No.13/212,036, now U.S. Pat. No. 8,238,318, and Ser. No. 13/536,927 andincorporated herein, modem 624 of FIGS. 6 and 7 produces K transmitsymbol streams wherein each of the K transmit symbol streams comprises asequence of blocks of modulated symbols. In a PTP configuration, the Ktransmit symbol streams would be destined to a peer receiver at theother IBR in the link. In a PMP configuration at the AE-IBR, one or moreof the K transmit symbol streams, as designated by the AE-IBR, would bedestined to the receiver in each one of the RE-IBRs. Also, in a PMPconfiguration at the RE-IBR, the K transmit symbol streams would bedestined to the receiver in the AE-IBR. Additionally as described ingreater detail in U.S. patent application Ser. No. 13/212,036, now U.S.Pat. No. 8,238,318, and Ser. No. 13/536,927 and incorporated herein,channel MUX 628 of FIGS. 6 and 7 generates M transmit chain inputsignals, wherein M≧K, and each of the M transmit chain input signals maybe generated with contribution from one or more (or all) of the Ktransmit symbol streams.

As described in greater detail in U.S. patent application Ser. No.13/212,036, now U.S. Pat. No. 8,238,318, and Ser. No. 13/536,927 andincorporated herein, each of the M transmit chain input signals isconverted to a transmit RF signal by respective ones of Tx1 . . . TxM636 in FIGS. 6 and 7. In a PTP configuration, the M transmit RF signalswould be directed via elements of the antenna array 648 as set by theRRC 660 to a peer receiver at the other IBR in the link. In a PMPconfiguration at the AE-IBR, one or more of the M transmit RF signals,as designated by the AE-IBR, would be directed via elements of theantenna array 648 as set by the RRC 660 to the receiver in each one ofthe RE-IBRs. Also, in a PNIP configuration at the RE-IBR, the M transmitRF signals would be directed via elements of the antenna array 648 asset by the RRC 660 to the receiver in the AE-IBR. In an embodiment ofthe IBR Antenna Array (648), M_(s) RF Transmit Reference Signals (680)are passed from the IBR Antenna Array (648) to the Canceller MUX (670)for use in cancellation processing and in some embodiments calibrationoperations. In some embodiments M_(s) will be equal to the number ofTransmit Antenna Elements Q_(T) in a one to one relationship.Additionally, in some embodiments M_(s) will be equal to the number ofRF transmit signals M in a one to one relationship. In embodiments whereQ_(T) is greater than M, the number of M_(s) RF Transmit ReferenceSignals may be equal to M, in which case the RF switch fabric (812)within the antenna array (648) may be utilized to select M_(s) RFtransmit reference signals (680) of the Q_(T) available Transmit AntennaReference Signals (805). In such embodiments, in which M_(s) is equal tothe number of RF Transmit Signals M, and in which the number of TransmitAntenna Elements (652) Q_(T) is greater than the number of RF TransmitSignals M, the selection of the specific M_(s) RF Transmit ReferenceSignals will be made by the RRC (660), and in correspondence with theselection of the Antenna Elements (652) utilized for the transmission ofthe RF Transmit Signals and further may utilize the same selectioncontrol signaling or alternative selection control signaling.

In alternative embodiments, the M_(s) RF Transmit Reference Signals(680) may be obtained directly from the M_(s) RF Transmit Signals withinCancellation MUX 670, rather than from the Antenna Array 648. In variousembodiments, the M_(s) RF Transmit Reference Signals (680) may beutilized in analog or digital cancellation processing as will bedescribed in further detail in relation to subsequent figures.

The receive signal processing path in FIGS. 6 and 7 of exemplary IBRs,except for the processes and structures associated with the CancellationMUX 670, conceptually reverses the operations performed in one or morepeer IBR transmitters providing signals to a particular IBR receiver. Asdescribed in greater detail in U.S. patent application Ser. No.13/212,036, now U.S. Pat. No. 8,238,318, and Ser. No. 13/536,927 andincorporated herein, N receive RF signals are provided by variouselements of the antenna array 648 as set by the RRC 660 and thenconverted to N receive chain output signals by respective ones of Rx1 .. . RxN 640 in FIGS. 6 and 7. Additionally as described in greaterdetail in U.S. patent application Ser. No. 13/212,036, now U.S. Pat. No.8,238,318, and Ser. No. 13/536,927 and incorporated herein, channel MUX628 of FIGS. 6 and 7 generates L receive symbol streams, wherein N≧L,and each of the L receive symbol streams may be generated withcontribution from one or more (or all) of the N receive chain outputsignals. As described in greater detail in U.S. patent application Ser.No. 13/212,036, now U.S. Pat. No. 8,238,318, and Ser. No. 13/536,927 andincorporated herein, modem 624 of FIGS. 6 and 7 demodulates one or more(or all) of the L receive symbol streams.

FIG. 8 illustrates an exemplary embodiment of an IBR Antenna Array 648with dedicated transmission and reception antennas. FIG. 8 illustratesan antenna array having Q_(R)+Q_(T) directive gain antennas 652 (i.e.,where the number of antennas is greater than 1). In FIG. 8, the IBRAntenna Array 648 includes an IBR RF Switch Fabric 812, RFinterconnections 804, a set of Front-ends 809 and 810 and the directivegain antennas 652. The RF interconnections 804 can be, for example,circuit board traces and/or coaxial cables. The RF interconnections 804connect the IBR RF Switch Fabric 812 and the set of Front-endTransmission Units 809 and the set of Front-end Reception Units 810.Each Front-end Transmission Unit 809 is associated with an individualdirective gain antenna 652, numbered consecutively from 1 to Q_(T).Additionally, the IBR RF Switch Fabric 812 is further coupled to receiveRF Transmit Reference Signals 805 from each Front-end Transmission unit,in specific embodiments, to allow for the selection M_(s) of the Q_(T)RF Transmit Antenna Signals, as previously described, to be provided tothe Cancellation Mux 670 as RF Transmit Reference Signals (1 . . .M_(s)) 680. Each Front-end Reception Unit 810 is associated with anindividual directive gain antenna 652, numbered consecutively from 1 toQ_(R). The present embodiment may be used, for example, with the antennaarray embodiments of FIGS. 11 and 12, or those depicted in U.S. patentapplication Ser. No. 13/212,036, now U.S. Pat. No. 8,238,318, and Ser.No. 13/536,927 and incorporated herein. Exemplary embodiments of the IBRRF Switch Fabric 812 are also described in detail in U.S. patentapplication Ser. No. 13/212,036, now U.S. Pat. No. 8,238,318, and Ser.No. 13/536,927 and incorporated herein. For example, in some embodimentsthe IBR RF Switch Fabric 812 provides the capability to connect any ofthe M transmit RF signals to any of the Q_(T) Front-end TransmissionUnits 809 with associated individual directive gain antenna 652, or toconnect any of the N receive RF signals to any of the Q_(R) Front-endReception Units 810 with associated individual directive gain antenna652.

In an alternative embodiment, the IBR RF Switch fabric 812 may bebypassed for the transmission signals when the number of dedicatedtransmission antennas and associated Front-end Transmission Units(Q_(T)) is equal to the number of transmit RF signals (e.g. Q_(T)=M),resulting in directly coupling the transmit RF signals from respectiveTx1 . . . TxM 636 to respective Front-end Transmission Units 809. In anassociated embodiment, the IBR RF switch fabric 812 may be bypassed forthe selection of the RF Transmit Reference Signals (680) coupled to theCancellation MUX (670), by directly connecting the RF Transmit ReferenceSignals (1 . . . Q_(T)) (805) directly to the RF Transmit ReferenceSignals (1 . . . M_(s)) (680), when M=Q_(T). In an additionalalternative embodiment, the IBR RF Switch fabric 812 may also bebypassed for the reception signals when the number of dedicatedreception antennas and associated Front-end Reception Units (Q_(R)) isequal to the number of receive RF signals (e.g. Q_(R)=N), resulting indirectly coupling the receive RF signals for respective Rx1 . . . RxN640 to respective Front-end Reception Units 810. Alternatively, the IBRRF Switch fabric 812 may also comprise circuitry to combine signals fromtwo or more Front-end Reception Units or to provide signals to two ormore Front-end Transmission Units as described in greater detail in U.S.patent application Ser. No. 13/212,036, now U.S. Pat. No. 8,238,318, andSer. No. 13/536,927 and incorporated herein.

As shown in FIGS. 9A and 9B, each Front-end 809 or 810 also includes an“Enable” input 925, 930 that causes substantially all active circuitryto power-down. Power-down techniques are well known. Power-down isadvantageous for IBRs in which not all of the antennas are utilized atall times. It will be appreciated that alternative embodiments of theIBR Antenna Array may not utilize the “Enable” input 925, 930 orpower-down feature. With respect to FIG. 9A, Bandpass filter 940receives transmission signal RF-SW-Tx-qt, provides filtering and couplesthe signal to power amplifier 904, then to low pass filter 950. Theoutput of the lowpass filter is then coupled to a dedicated transmissionantenna, which is comprised of directive antenna element 652 with gainGqt. FIG. 9A also depicts the RF Transmit Reference Signal (805) whichin this exemplary embodiment may be obtained by a line coupler (955) tothe interconnection between low pass filter 950 and directive antennaelement 652 wherein the utilization of such RF Transmit Reference Signal(805) within the ZDD Canceller 670 is described in greater detailherein. With respect to FIG. 9B, directive antenna element 652 with gainGqr is a dedicated receive only antenna and coupled to receive filter970, which is in turn coupled to LNA 908. The resulting amplifiedreceive signal is coupled to band bass filter 960, which provides outputRF-SW-Rx-qr.

FIG. 10 is a diagram of an exemplary horizontally arranged intelligentbackhaul radio antenna array intended for operation in the 5 to 6 GHzband and FIG. 11 is a diagram of an exemplary vertically arrangedintelligent backhaul radio antenna array also intended for operation inthe 5 to 6 GHz band. Analogous versions of the arrangement shown inFIGS. 10 and 11 are possible for any bands within the range of at least500 MHz to 100 GHz as will be appreciated by those of skill in the artof antenna design. In both FIG. 10 and lithe number of transmitdirective antenna elements 652 and associated Front-end TransmissionUnits 809 is Q_(T)=2. Larger values of Q_(T) are straightforward toimplement by increasing the width of the antenna array depicted in FIG.10 or possibly without increasing any outside dimensions of the antennaarray depicted in FIG. 11 at least for Q_(T)=4 as will be appreciated bythose of skill in the art of antenna design. In both FIG. 10 and lithenumber of receive directive antenna elements 652 and associatedFront-end Reception Units 810 is Q_(R)=8. Larger values of Q_(R) arestraightforward to implement by increasing the width of the antennaarray depicted in FIG. 10 or by increasing the width and/or height ofthe antenna array depicted in FIG. 11 as will be appreciated by those ofskill in the art of antenna design.

The transmit directive antenna elements depicted in FIGS. 10 and 11comprise multiple dipole radiators arranged for either dual slant 45degree polarization (FIG. 10) or dual vertical and horizontalpolarization (FIG. 11) with elevation array gain as described in greaterdetail in U.S. patent application Ser. No. 13/536,927 and incorporatedherein. In one exemplary embodiment, each transmit directive antennaelement has an azimuthal beam width of approximately 100-120 degrees andan elevation beam width of approximately 15 degrees for a gain Gqt ofapproximately 12 dB.

The receive directive antenna elements depicted in FIGS. 10 and 11comprise multiple patch radiators arranged for either dual slant 45degree polarization (FIG. 10) or dual vertical and horizontalpolarization (FIG. 11) with elevation array gain and azimuthal arraygain as described in greater detail in U.S. patent application Ser. No.13/536,927 and incorporated herein. In one exemplary embodiment, eachreceive directive antenna element has an azimuthal beam width ofapproximately 40 degrees and an elevation beam width of approximately 15degrees for a gain Gqr of approximately 16 dB.

Other directive antenna element types are also known to those of skillin the art of antenna design including certain types described ingreater detail in U.S. patent application Ser. No. 13/536,927 andincorporated herein.

Preliminary measurements of exemplary antenna arrays similar to thosedepicted in FIG. 10 show isolation of approximately 40 to 50 dB betweenindividual transmit directive antenna elements and individual receivedirective antenna elements of same polarization with an exemplarycircuit board and metallic case behind the radiating elements and aplastic radome in front of the radiating elements. Analogous preliminarymeasurements of exemplary antenna arrays similar to those depicted inFIG. 11 show possible isolation improvements of up to 10 to 20 dB forsimilar directive gain elements relative to FIG. 10. Thus, for certainIBR embodiments in ZDD operation, the vertical antenna array arrangementdepicted in FIG. 11 may be preferable to the horizontal antenna arrayarrangement depicted in FIG. 10, providing for additional initial RFisolation.

FIG. 12 is a block diagram of a portion of an IBR according to oneembodiment of the invention that illustrates one exemplary embodiment ofthe ZDD Canceller 670 in greater detail internally and with relationshipto other parts of an exemplary IBR. The implementation philosophy shownin the current embodiment is based on cancelling the transmit chainsignals from the IBR Transmit Antenna Array (648A) that undesirably leakin to the receive chains (comprised of signal flow from IBR ReceiveAntenna Array (648B) to the receive portion of the IBR ChannelMultiplexer (628)). ZDD cancellation may be performed using multipleapproaches and at various stages within the IBR receive chains. Withinalternative embodiments, it is also possible to cancel the transmitstreams (1 . . . K, from IBR Modem 624) within the receive streams (1 .. . L, output from the IBR Channel Multiplexer 628). Further alternativeembodiments employ cancellation of the transmit chain signals within thereceive streams (1 . . . L, output from the IBR Channel Multiplexer628).

Note that within the current embodiment, associated with performingcancellation at various stages of the receiver chains, cancellation maybe performed at analog baseband, intermediate frequency (IF), RF and/ordigital baseband.

In order to achieve the required performance of the IBR 1200, the signalto noise ratio of the Receive Streams (1 . . . L) must be sufficient soas to allow for acceptable demodulation error rate at IBR Modem (624).As discussed above, conventional radios utilize frequency duplexing ortime duplexing to allow for sufficient isolation of the transmittersignals from the signal being received and demodulated. Associated withthe exemplary embodiments of the ZDD enabled IBR 1200, isolation of thedesired receive signals from transmitted signals is accomplishedutilizing a combination of active cancellation and inherent isolationbetween the IBR Transmit Antenna Array (648A), and the IBR ReceiveAntenna Array (648B). The various isolating features and functions arereferred to in the following discussion as Isolation Loops orCancellation Loops. Embodiments of the Cancellation Loops generallyinclude adaptation based on active measurements of signals, channelestimates, cancellation metrics, or other metrics. In the currentembodiment depicted in FIG. 12, four loops are shown: Loop L0 (1202), L1(1204), L2 (1206), and L3 (1208). Note each individual loop may becomprised of multiple successive or nested loops of similar orequivalent function, collectively operating as a single loop.

The Isolation “Loop” L0 (1202) is mainly just indicative of the finiteisolation between any two antennas (Tx to Rx, or Tx to Tx) which is acritical parameter for FDD and even more critical in ZDD. Someembodiments of L0 (1202) will not include adaptive adjustment or utilizeactive control, but may still be referred to as a Loop for the constancyof the terminology herein. Other embodiments of L0, are truly a “loop”to the extent that some feedback mechanism either moves a servo to anisolating structure or a tuning element that affects a differentisolation transfer function between any two antennas or sets of antennasof interest for a particular operational mode. Such tuning of controlmay further include the optional antenna selection function of RxAntenna/Channel Switch Matrix within 648B. The operation of the optionalRx Antenna/Channel Switch Matrix within 648B is equivalent to that ofthe receive portions of IBR RF Switch Fabric 812 of FIG. 8, wherein theselection is performed of N RF Receive Signals (RF-Rx-1 to RF-Rx-N) ofthe Q_(R) available RF Receive Antenna Signals (labeled consecutivelyY₁(t) to Y_(QR)(t)) shown in FIG. 12. Such RF Receive Signals (678) areprovided to canceller 670, and more specifically to canceller RF summingnodes (alternatively referred to as RF cancellation combiners) 1210-1 to1210-N respectively.

Design of the antenna array for minimal magnitude response by and/orcapability of L0 tuning by feedback is a key enabler for simplifying thedemands of L1, L2 and/or L3. Relative to L0 (1202), it is generallyexpected, and experimentally confirmed in specific embodiments allowingfor testing, IBR antenna elements and orientations in a “vertical” arraystack as shown in FIG. 11 will have better isolation due to the higherelevation gain than azimuthal gain amongst the elements, relative to thehorizontal configurations depicted in FIG. 10.

The Canceller Loop L1 (1204) samples the “actual” transmitted RF signalin each Tx chain (RF Transmit Reference Signals 1 to M_(s) (680) labeled680-1 to 680-M), and then generates for each Rx chain a modified“cancellation signal C1 _(n)” (also referred to in some embodiments as aRF transmit leakage cancellation signal) that when summed (via RFsummers 1210-1 to 1210-N) with the Rx chain signals (678) before inputto the downconverters (640) substantially cancels the Tx signals 1through M (X₁(t) to X_(M)(t)) that have leaked into the Rx antennascontributing to each Rx chain “n”. Exemplary embodiments of L1 (1204)are realized by an analog equivalent to a complex FIR “filter”implemented at RF and described in relation to FIG. 13. In alternativeimplementations L1 (1204) may be implemented by down converting RFTransmit Reference Signals 680 to an intermediate frequency (IF) for FIRprocessing and then either upconverted back to RF or applied at acancellation-summing node at IF.

For some embodiments of ZDD-IBRs, L1 may be targeted at only the largestTx to Rx coupling paths, and those that are fixed for a specific IBR RFswitch fabric (812) selection. Such coupling paths are expected toinvolve timescales with variations typically of order 1 ns or less forexample in some embodiments. In specific embodiments it is possible todetermine the appropriate loop coefficients once (at a factorycalibration in some embodiments) and then refine only very occasionally.Such an approach may be utilized in other embodiments where longer timedelays with mode variation are addressed as well, as a combination of afixed or slowly adapting L1, and a parallel of sequential L1 cancellerswhich address the longer delay, lower magnitude, and more highly varyingcoupling paths. Such embodiments may include a “primary” loop to addressthe most significant magnitude response components and once cancelled bythe primary loop (typically including L1 but possibly using part of L2or L3 instead or in addition to L1), this should allow a secondary loop(probably within L2 or L3) to track shorter variations of much smallermagnitude. Details of exemplary L1 embodiments are described in furtherdetail in connection with FIG. 13.

The Canceller Loop L2 (1206) also samples the transmitted RF signals (RFTransmit Reference signals 680). In some embodiments, it may bepreferable not to sample directly as indicated but instead sample bytaking a set of interim output signals from within L1 (there are M×Nsuch interim cancellation signals) and then applying the additional fineresolution processing of L2 (1206). L2 is distinctly different from L1(1204) in that L2 processes Tx signals in the digital baseband domainusing FIR digital “filter” techniques which enables L2 to practicallycancel far lower power signals with substantially longer delays thanwith L1 (1204). Optionally the L2 cancellation signal can be upconvertedback to RF (“C2R_(n)” as shown) or used as a cancellation signal atdigital baseband (“C2D_(n)” as shown and referred to as a basebandtransmit leakage cancellation signal in some embodiments). Inembodiments utilizing digital baseband cancellation via L2 or L2 and L3together, summer nodes (alternatively referred to as cancellationcombiners) 1212-n will be respectively utilized to sum the C2D_(n)signal with receive chain output signals (674) from respective receivechains to provide respective baseband cancelled receive signals. A majorissue With L2 is the noise and distortion added by the act first ofdownconverting and digitizing the sampled Tx RF signals, and thenfurther upconverting and leveling if the optional “C2R_(n)” signals areto be generated. Aspects of L2 embodiments addressing such impairmentswill be discussed. Note that exemplary embodiments of such processingare depicted and described in further detail in connection with FIG. 14.

The Canceller Loop L3 (1208) is very different from L1 (1204) or L2(1206) in that it takes as an input a digital baseband representation ofeach Tx chain signal Tx₁(t) to Tx_(M)(t). In L3 (1208) all processing ofthe Tx chain signal can be done using digital FIR “filter” techniquessimilar to L2. Ideally the input to L3 would be after the digital lowpass channel filter 1214-m. In embodiments where the filter outputs arenot accessible the digital filters (1214) can also be replicated in L3to generate a better estimate of the actual Tx signal. Such replicationprovides for matched channel impacts thus reducing the overall need forthe L3 channel estimation to include compensation for filters 1214-1 to1214-M. Eliminating the requirement to estimate this filter will allowfor a faster or less complex estimation of L3 cancellation coefficientsin some embodiments. Similarly, the effective response of the low passanalog filter (1216-m) following the DAC, and the BPF (1218-m) followingthe upconverter can also be included in L3 to improve the cancellationaccuracy, efficiency, or convergence speed. Additionally, it iscontemplated that other filters may be included in L3 as well, forinstance those in the receiver such as a bandpass filter (not show)following summers 1210-n, and lowpass filters 1221-n, and 1222-n.Further, certain intermod products in the Tx chain or as created in theRx front-ends (or in the analog portions of L1 or L2) can also bemodeled within L3 (or possibly within L2) for cancellation by the L3output. Such processing will be further described in connection withembodiments depicted in FIG. 15.

However, L3 (1208) cannot cancel Tx chain noise as L1 can generally orL2 can subject to certain limitations. Typically the L3 cancellationsignals “C3D_(n)” would be used at digital baseband (referred to asbaseband transmit leakage cancellation signals in some embodiments) asshown but optionally these signals can be upconverted as “C3R_(n)” andapplied at RF (at summer 1210-n for instance and wherein C3R_(n)comprises an up-converted baseband transmit leakage cancellation signalwhich acts as a digitally generated RF transmit leakage cancellationsignal in some embodiments). It is also possible to generate both“C3D_(n)” and “C3R_(n)” simultaneously and in some embodiments withdifferent emphases on the various undesired signal components to becancelled.

The ZDD Canceller Loop Coefficients Generator (ZCLCG) (675) is notionalas shown in FIG. 12. This illustrates a bus for coefficients to betransferred to the various loops (and in some embodiments, operationinformation to be fed back from the various loops to the ZCLCG). It isshown with the primary inputs being the pass through digital basebandreceive chain signals from which the ZCLCG can measure the effects ofdifferent loop coefficients on the undesired signal levels present, aswell as perform calibration and channel estimation functions associatedwith the transmitter, receiver, and loop signal paths. However, otherinputs such as RSSI from each downconverter chain, or FFT data (orderived FFT data) for each Rx chain from the IBR channel multiplexer(not specifically illustrated above) may also be used by the ZCLCG todetermine various loop coefficients in specific embodiments.Furthermore, the ZCLCG itself as a “processor” may be within the sameresources as the IBR channel multiplexer and/or IBR modem or astandalone processor or a custom chip with one or more of the loops, orsome hybrid of some or all of the above in various embodiments.Embodiments of the ZCLCG will also include state machine functionalityso as to perform the aforementioned procedures as part of a process, orin reaction to events, or time durations and periods. For instance, inone embodiment, L0 may be modified based upon a change in the selectedreceive antennas to be utilized within the Receive Antenna Array (648B)and the IBR RF Switch Fabric (812). In response to this change, the loopcoefficients may be required to be modified by the ZCLCG (675). In oneembodiment, the timing of such an antenna selection change may be incoordination with one or more of the RRC (660), IBR MAC (612), RLC(656), and other functional blocks of FIGS. 6 and 7 including IBMS Agent(700), so as to cause the antenna selection change to be just prior to aknown signal transmission period designed and utilized for the trainingof the ZCLCG. Such an approach has the benefit of potentially minimizingthe loss of transmitted information to an intended receiving IBR, andmay further include pre-determined waveform properties beneficial to thetraining and loop coefficient determination. Other embodiments mayperform such a training process on a periodic basis, and/or in responseto a measured cancellation performance threshold, a temperature change,or changes in other IBR operating parameters. Embodiments may alsoutilize a training process wherein various loops are trainedsequentially and utilizing various procedures and algorithms. Forexample in one embodiment, upon a change to L0 (1202), L1 (1204) may betrained with L2 (1206) and L3 (1208) effectively turned off, andutilizing a closed loop adaptive algorithm with the goal of minimizing aparameter associated with a transmitted training signal (such as RSSI,or correlated signal power). Following the completion of the L1coefficients being set, the coefficients of successive loops may beprogressively determined and set. In some embodiments, the determinationof the coefficients may be performed adaptively using a steepest decentclass of algorithm, while in other embodiments a close form calculationmay be utilized wherein a measurement of the impulse response orfrequency response (referred to herein generically as the channelresponse) is made including an actively cancelling L1. The channelresponse measurement would be made from each transmit chain to eachreceive chain in some embodiments, and result in M×N channelmeasurements (where M is the number of transmit chains, and N is thenumber of receive chains). Such a channel measurement may then beutilized to calculate the required loop coefficients for L2, if present,and L3. Such calculations may be performed with the same or separatechannel measurements for L2 and L3. In embodiments where separatechannel measurements are utilized for L2 and L3 coefficient calculation,some embodiments may perform the L3 channel measurements with L1 and L2cancellation active such that the cancellation performance is include inthe resulting L3 channel measurements.

Referring to the IBR RF (632) and specifically the receive portions(640), the receive gain control, in some specific embodiments, must takeinto account the remaining transmit signal following the L1 cancellationprocess at the summers 1210-n. Some embodiments may control the AGC fromthe ZDD Canceller Loop Coefficients Generator (675) or from the IBRModem or IBR Channel Mux instead of from within the Rx chain as shown inFIG. 12. Within the transmitter chains, transmit power control isexpected but not shown, and in some embodiments would be under thecontrol (at least partially) of the coefficient generator (675) so thecancellation performance may aid in the determination of a maximum ordesirable transmit power level based upon limitations of thecancellation multiplexer (670).

IBR embodiments utilizing ZDD may be utilized in at least two variants.The first variant is “co-channel” ZDD (“CC-ZDD”) wherein the Tx chainchannels at least partially overlap (if not completely coincide with)the Rx chain channels. Embodiments utilizing CC-ZDD are theoreticallypossible of achieving twice the spectral efficiency for a single link atthe physical layer, relative to systems not utilizing ZDD approaches.When coupled with available MAC efficiencies due to the reduction ofdelay and increased automatic retransmission (ARQ) efficiency utilizingan ACK/NAK protocol or the equivalent, additional efficiencies arepossible. In particular, the delay reduction relative the TDD basedsystems is significant. This CC-ZDD mode is ideal for licensed bandoperation where interference is well controlled and spectral efficiencymost highly valued.

The second ZDD variant can be called “single band” or “co-band” ZDD(CB-ZDD) wherein both the Tx chain channels and the Rx chain channelsare within a single band (using a single band pass filter) utilizedduring both transmission and reception, but the channels do not overlap.Such an arrangement results in minimal spectral efficiency improvement(though FDD is typically better than TDD) but is still highly desirableat least as a fallback mode for unlicensed band operation. One advantageof CB-ZDD in unlicensed bands results from the interference seen at thereceiver of one IBR, being dramatically different (and frequencydependent) compared to its peer device such that operating in a similarfrequency range allows for a more optimized channel frequency choiceunder significant interference conditions. Other advantages include theflexibility for a CB-ZDD device to operate in bands which operationwould otherwise not be possible without expensive and fixed band passfilters. In some ways, “co-band” ZDD (“CB-ZDD”) is simpler because theanalog and/or digital baseband low pass filters (as well as the FFTs inthe IBR channel multiplexer) can perform much of the Tx signalcancellation. But in other ways since the requirement to cancel the Txchain noise within the Rx chain channel remains the same as CC-ZDD, thenCB-ZDD has the additional complication that the “easily” detectable Txsignal that drives the ZCLCG process is not at the right channelfrequency to ensure noise cancellation in the Rx chain under normaloperating conditions. However other metrics for Tx noise cancellationmay be utilized mitigating much of this disadvantage, such as adaptingthe loops based upon receiver performance metrics such as signal tonoise ratio (SNR), bit error rate (BER), frame error rate (FER) ormetrics associated with the forward error correction (FEC). Suchapproaches would be most applicable to L1 or L2 with C2D_(n) adaptation(or “L2D”) for noise cancellation.

The aforementioned CC-ZDD (and potentially CB-ZDD) has specificapplicability in use in an ZDD Aggregation End IBR (AE-IBR) whenoperating in a point to multi-point (PMP) mode in communication withmultiple non-ZDD Remote End IBRs (RE-IBRs) operating each in a TDD mode.Such a configuration allows for the ZDD enabled AE-IBR to betransmitting to one or more RE-MRs, whilst receiving from one or moreother RE-IBRs. In these embodiments, no ZDD cancellation is required atthe RE-IBRs, but a doubling of the overall network efficiency isrealized relative to AE-IBRs not utilizing ZDD. To enable such anembodiment, a time multiplexing of one or more of the AE-RE links andRE-AE links must be arranged and scheduled such that the RE-IBRs aretime multiplexed with their transmission and reception periods, andoffset relative to other RE-IBRs, at least to the extent that TDDoperation is achievable at each RE-IBR individually, not consideringother multipoint multiplexing approaches such as frequency multiplexing(such as Orthogonal Frequency Division Multiple Access (OFDMA), SingleCarrier Frequency Division Multiplexing (SC-FDM), and the like). Suchtechniques may be using in combination with the aforementioned ZDDmultiplexing approaches.

Other embodiments of the aforementioned ZDD techniques may be utilizedfor repeaters to be used interposed between IBRs, or in conjunction as afeature of a particular ZDD enabled IBR. Such repeater embodiments mayutilize ZDD approaches to allow for reception and transmission of signalsimultaneously allowing for higher efficiency relative to TDD basedrepeating approaches, or more spectral efficiency relative to FDD basedrepeating approaches. ZDD repeater embodiments may perform the repeatingfunction in a number of approaches including: at an RF andun-demodulated level, a modulation symbol by symbol level, a streamlevel, an FEC Block Level, a MAC Frame Level, or potentially higherlevels. When performed at a stream level, beam forming techniques may beutilized to allow for a spatially rich propagation environment allowingfor an increased performance network.

Embodiments of a Loop 1 (L1) (1204) canceller of cancellationmultiplexer (670) are depicted in FIG. 13. An exemplary Loop L1embodiment samples the “actual” transmitted RF signal in each Tx chain(1302-m), and then generates for each Rx chain a modified “cancellationsignal C1 _(n)” (1318-n) that when applied to summing node (1210-n) withthe Rx chain signal (678-n) before input to the downconverter (1220-n)substantially cancels the Tx signals 1 through M that have leaked intothe Rx antennas contributing to Rx chain “n”. Some embodiments of L1 arerealized by an analog equivalent to a complex FIR “filter” implementedat RF. Alternative embodiments of L1 provide for signal to be downconverted to an IF for FIR processing and then either up converted backto RF or applied at IF.

The transfer function G_(Sm) (1306-m) is intended to indicate a“zero-th” order match between the magnitude and delay from X_(m)(t) toan average Rx chain at the first summer and a known frequency response.The delay might be realized in a cable of preset length thatapproximates the delays in the Tx-Rx path. The transfer function mightbe strongly influenced by one or more band pass filters in each Rxchain, or other filters in either the transmit or receive paths whichare “in the loop”, and so it may be advantageous to place an identicalband pass filter (or other such “in the loop” filters) within eachG_(Sm) (1306-m). This has the additional benefit of keeping out of bandTx spurs from being injected into the Rx chains. The transfer functionalso includes the effect of the coupling to X_(m)(t), by a line couplernear the Tx antenna feed point in some embodiments. It is desirable tosample a signal highly correlated to the actual X_(m)(t) transmissionand to have sufficient signal such that the sampled noise floor is farabove the equivalent input noise of L1.

In some embodiments of the “FIR” structure depicted in FIG. 13, a delayline (1310) is of length D1*Δt₁ and for each delay tap, weightsW1_(m,n,d) (1311) are complex with separate I and Q components, while inother embodiments such weights may be in amplitude only. In an exemplaryembodiment as depicted by FIG. 13, the weights (1311) are typically sentfrom the ZCLCG (675) as finite bit words (i.e. 8, 10 or 12 bits per I orQ component). In an alternative embodiment, the weights (1311) come froman analog feedback loop. In an exemplary embodiment as depicted by FIG.13, each weight W1_(m,n,d) (1311) is complex multiplied (1312) in the RFsignal domain on the d-th delayed version of the m-th sampled Tx chainsignal and then the results are summed across each d=0 to D1 for eachmat summing nodes 1314, and then across all m=1 to M for each of n=1 toN at summing nodes 1316-n.

The exemplary embodiment depicted in FIG. 13 indicates a delay line(1310) of uniform steps Δt₁ from d=0 to d=D1 (similar to a FIR filterstructure). However, it may be neither practical nor desirable to haveuniform steps. If L1 is “matched” to reflections including transmit toreceive propagation time delays up to a maximum time delay (such amaximum time delay sufficient to address all reflections above aspecific magnitude threshold) then there may be non-uniform time stepsthat result in a better cancellation by design. Also, it may be neitherpractical nor desirable to have uniform amplitudes from each tap of thedelay line for analogous reasons. In some embodiments the typical delaywithin G_(Sm) will be of order 1 ns, and the delay line length will beof similar order or possibly less. In a discrete implementation even 3“taps” may be impractical for a 2×4 system, but when implemented in acustom RFIC it may be practical to have on the order of 5-10 delaysteps, while other embodiments may have yet more taps.

In practice, the RF delay line (1310) depicted in FIG. 13 can berealized from a distributed transmission line with multiple taps, orfrom a lumped circuit equivalent with passive inductors and capacitors.In embodiments utilizing complex multipliers (1312) or equivalentcircuits thereof, each complex multiplier (1312) can be constructed, forexample only, from a pair of four-quadrant multipliers wherein onemultiplier is driven by an “I” set of weights and the other by a “Q” setof weights (collectively a complex weight 1310 in the currentembodiment), and further wherein either the RF input signal to thesecond multiplier is shifted 90 degrees relative to the first or theoutputs of the respective multipliers are summed (at summer 1314 in someembodiments) in quadrature instead of linearly. There are numerousconventional techniques to realize the RF summations (1314, 1316-n) inFIG. 13 and to combine one or more of summing nodes 1314, 1316-n and/or1210-n into a single summing node (wherein each summer may be referredto as a combiner elsewhere). It is also possible to implement thecanceller L1 without using quadrature signal paths and weights relyingon “real” sampling theory, by ensuring the taps are spaced to satisfythe Nyquist band pass sampling theorem. One embodiment of a delay lineutilizing complex samples and weights may be implemented with a singledelay line and linear summation, wherein each I and Q sample in thedelay line are sampled from different taps spaced at 1/(fo*4) in time,where fo is the center frequency of operation of the band, and if thedelay between each “pair” of taps is equated to 1/BW, then BW is thefrequency bandwidth of the cancellation bandwidth. In such anembodiment, each pair of taps, comprise in phase and quadrature phase(or real and imaginary) samples at a specific “tap”.

Referring now to FIG. 14, a block diagram of a Loop 2 ZDD cancelleraccording to one embodiment of the invention is depicted. As previouslydiscussed, the Canceller Loop L2 (1206) also samples the transmitted RFsignals though in some implementations it may be preferable not tosample directly as indicated but instead by taking a set of interimoutput signals from within L1 (1204) as there are M×N such interimcancellation signals in some embodiments and then applying theadditional fine resolution processing of L2. In other embodiments, the“FIR” filter structures of L1 may be incorporated into L2 (1206). Suchan arrangement may be utilized in embodiments not having a L1 (1204),but utilizing advantages of combining transversal filtering at RF ofeach of the transmitters individually (utilizing L1 weights) into all ofthe sampling receivers of L2. Such an arrangement allows for advantagesin capturing and utilizing transmitter and receiver noise (especiallyphase noise) in specific combinations. In such an embodiment it ispossible to capture all (M times N) combinations of the transmitter andreceiver multiplicative phase noises. In one embodiment of such anarrangement, passive components and micro-strip lines utilize a printedcircuit board or other substrate. Such arrangements allow for a “spatialmultiplexing” by the RF filter structures of all transmitter signalsonto each of the sampling receivers. Further embodiments allow for useof such FIR structures, or delay line structures to spatial multiplex RFTransmit Reference Signals (680-m) with receive chain signals (678-n) byutilizing either the signal receive chains (1225-n) and/or the samplingreceive chains (1406-m). In the case where both L2 sampling receivechains (1406-m) and signal receive chains (1225-n) are utilized, receivechain signals 678-n and RF Transmit Reference Signals (680-m) would becoupled to more than one of the utilized signal receiver chains (1225-n)and sampling receive chains (1406-m) to multiplex signals. Furtheradditional processing to de-multiplex the signals will be performed tode-multiplex the receive chain and sampling receive chain signals priorto applications to the L2 FIR filters (1408). Other embodiments mayutilize arrangements of signal receive chains (1225-n) alone or samplingreceive chains (alternatively referred to as transmit RF referencereceive chains in some embodiments) (1406-m) alone. Baseband processingequivalent to MIMO receiver processing may be utilized to separate theindividual combinations of transmitter to sampling receiver signals.When specific arrangements of shared VCOs are utilized among receiversand sampling receivers and between transmitters and in some cases C3R upconverter chains, the VCO phase noise impact may be greatly mitigated oreven eliminated.

In one greatly simplified and exemplary embodiment addressing theimpacts of phase noise (and other sources of noise), a single shared VCOis utilized for all up converters and down converters for thetransmitters, receivers, sampling receivers, and any C3R up converters.Such an arrangement does not require the RF “spatial multiplexing”discussed above in order to address phase noise concerns as thetransmitter phase noise when received using a C2D sampling receiver(1406, alternatively referred to as transmit RF reference receive chainsin some embodiments) will have a common noise as the transmitter signalwhen received on the signal receivers (1225-1 through 1225-N). Otherarrangements may share VCOs between all up converters (that generateC3R_(n)) and transmit chains (636), and between all down converterswithin C2D sampling receivers (1406-m of 1404-1 through 1404-M, andalternatively referred to as transmit RF reference receive chains insome embodiments) and signal receivers (1225-1 through 1225-N). Furtherembodiments may utilize common VCOs between pairs of down converters ina respective sampling receiver and signal receiver, or similarly utilizecommon VCOs for pairs of up converters for a respective signaltransmitters and C3R up converter chain.

Returning now to the exemplary embodiments of FIG. 14, embodiments of L2are distinctly different from L1 in that L2 processes Tx signals in thedigital baseband domain using FIR “filter” (1408) techniques that enableL2 to cancel far lower magnitude signals with substantially longerdelays than with L1. In some embodiments the L2 cancellation signal canbe upconverted back to RF (“C2R_(n)” not shown in FIG. 14) or used as acancellation signal at digital baseband (“C2Dn” as shown). One majorissue with L2 is the noise and distortion added by the act first ofdownconverting and digitizing the sampled Tx RF signals (1406).

In some embodiments, the inputs from X_(m)(t) can be a line coupledinput as described for L1, or can be from the same G_(Sm) (1306-m) (or aparallel G_(Sm) for L2 only) as described for L1. In alternativeembodiments, the inputs for each X_(m)(t) might be derived from the setof C1_(m,n) (where n=1 to N). This could be the average of the Ncancellation signals, the maximum of them or some weighted blend wherethe weights are determined dynamically by the ZCLCG.

Note that with respect to embodiments of FIG. 14, the number oftransmitters M may be different from the number of receivers N. Howeverin specific embodiments discussed above addressing the “spatialmultiplexing” of transmit reference signals (680-1 through 680-M) ontothe sampling receivers (1406-m of 1404-1 through 1404-M)) andpotentially the receive chains, the number of transmitters and receiversare equal (M=N).

In one embodiment in which C1_(n) (1318-n) signals are utilizedrespectively as inputs (1402-1 to 1402-M, where M=N) to each samplingreceiver 1406-1 to 1406-M, sets of weights (1311) may be chosen so as toallow for a “spatial” multiplexing between the transmitter signals andthe separation of each of the transmitter signal components from oneanother including associated transmitter noise and specific samplingreceiver (1406-m) noise as discussed above. As noted previously thephase noise of both the Tx chains and the Rx chains are particularlyproblematic for the operation of L2D processing. However in specificimplementations, utilizing transmitters and receivers each with two ormore shared voltage controlled oscillators and frequency references,such noise impact may be compensated for in such embodiments. In such aconfiguration, all M transmitters will be received by all N samplingreceivers (1406-1 to 1406-M, when M=N). As noted, by appropriatelychoosing the L1 weights (1311) associated with each RF transmitterreference signal 1302-m, each combination (M by N) of transmitter andreceiver noises will be sampled and recoverable via the mentionedspatial multiplexing approach. To the extent that pairs (or more) ofreceivers have common frequency local oscillators (VCOs and frequencyreferences) individual transmitter and receiver noise components may berecovered and or be compensated for so as to allow for base band digitalcancellation of the noise included in the Tx to Rx leakage signals fromeach receive chain. This is enabled by pairing each signal receiver(1225-n) with an L2D sampling receiver (1406-m) with a common L0frequency as discussed. The demultiplexing may be performed utilizingspecific weights and processing within the FIR filters of FIG. 14 (1408for example). In one example the so called filter weights for each FIRfilter 1408 may be comprised of the convolution of weights forcoefficients to separate specific transmitter and receiver components,and to compensate for Tx to Rx frequency responses to allow forcancellation. Alternatively such processing may be performed in thefrequency domain rather that in the time domain or in additionalprocessing not depicted in FIG. 14.

In the embodiments associated with FIG. 14, all sets of weights (1407)are complex sets of d=1 to D2 individual tap weights each typically withseparate I and Q components. They are typically sent from the ZCLCG(675) as finite bit words (i.e. 12, 16, or more bits per I or Qcomponent). Each individual tap weight W2_(m,n,d) (1407) is complexmultiplied digitally on the d-th delayed version of the m-th sampled Txchain signal and then the results are summed across each d=0 to D2 foreach m (within each FIR2_(m,n) 1408), and then across all m=1 to M (ateach summer 1410-n) for each of n=1 to N.

The preceding embodiment indicates a conventional complex FIR filterstructure for CFIR2 (1408). It may be desirable when L2 is being used tocancel significant variations in overall delay of the Tx to Rx chaincoupling paths to, in some embodiments, have a complex CFIR2 with asmaller time delay for certain of the taps and longer for the remainder,or to have parallel complex FIR structures with different time delaysteps and then sum them together.

In some embodiments each sampling receiver chain (1406-m) may berealized in practice using conventional components or incorporated in anRFIC. Each CFIR2_(m,n) (1408) and digital domain summation (1410-n) canbe constructed using conventional digital circuits in either an ASIC orFPGA, or realized in software on a DSP.

Referring to FIG. 15, a block diagram of embodiments of a Loop 3 ZDDcanceller is depicted. The Canceller Loop L3 (1208) is very differentfrom L1 (1204) or L2 (1206) in that it takes as an input a digitalbaseband representation of each Tx chain signal Tx₁(t) to Tx_(M)(t). InL3 (1208) all processing of the Tx chain signal can be done usingdigital FIR “filter” techniques similar to L2. Ideally the input to L3would be after the digital low pass channel filter 1214-m. Inembodiments where the filter outputs are not accessible the digitalfilters (1214) can also be replicated in L3 to generate a betterestimate of the actual Tx signal. Such replication provides for matchedchannel impacts thus reducing the overall need for the L3 channelestimation to include compensation for filters 1214-1 to 1214-M.Eliminating the requirement to estimate this filter will allow of afaster or less complex estimation of L3 cancellation coefficients insome embodiments. Similarly, the effective response of the low passanalog filter (1216-m) following the DAC, and the BPF (1218-m) followingthe upconverter can also be included in L3 to improve the cancellationaccuracy, efficiency, or convergence speed. Additionally, it iscontemplated that other filters may be included in L3 as well, forinstance those in the receiver such as a bandpass filter (not show)following summers (alternatively referred to as cancellation combiners)1210-n, and lowpass filters 1221-n, and 1222-n. Further, certainintermod products in the Tx chain or as created in the Rx front-ends (orin the analog portions of L1 or L2) can also be modeled within L3 (orpossibly within L2) for cancellation by the L3 output.

However, L3 (1208) cannot cancel Tx chain noise as L1 can generally orL2 can subject to certain limitations. Techniques to mitigate such noiseimpacts may be addressed associated with L1 or L2 as discussed. One suchapproach combining C2D_(n) and C3R_(n) will be discussed associated withFIG. 18. In particular, an approach for capturing correlated C3R_(n)noise components by a C2D_(n) receiver for later cancellation has beendiscussed and is disclosed in greater detail.

Returning to the exemplary embodiment of FIG. 15, the L3 cancellationsignals “C3D_(n)”, where n is from 1 to N, would be used at digitalbaseband as shown but optionally these signals can be upconverted as“C3R_(n)” and applied at RF. It is also possible to generate both“C3D_(n)” and “C3R_(n)” simultaneously and possibly with differentemphases on the various undesired signal components to be cancelled,however this would require additional FIR circuitry (not shown) toaccount for the different magnitude, phase and delay between C3R_(n) andC3D_(n) even for the “same” cancellation signal effect.

In the exemplary embodiment of FIG. 15, all weights W3_(m,n) are complexsets of d=1 to D3 individual tap weights each typically with separate Iand Q components and utilized within CFIR3_(m,n) (1516-n within 1508-m).In specific embodiments weights are typically sent from the ZCLCG (675)as finite bit words (i.e. 12, 16, or more bits per I or Q component).Each individual tap weight W3_(m,n,d) is complex multiplied digitally onthe d-th delayed version of the m-th sampled Tx chain signal and thenthe results are summed across d=0 to D3 for each m within each CFIR3(1516-n within 1508-m), and then across all m=1 to M (at summer 1520-n)for each of n=1 to N.

The above discussion indicates the use of a conventional FIR filterstructure for CFIR3 (1516-n for example). It may be desirable when L3 isbeing used to cancel signal components with significant time delayvariations to either have a complex CFIR3 with a smaller time delay forcertain of the taps and longer for the remainder, or to have parallelcomplex FIR structures with different time delay steps and then sum themtogether.

The exemplary embodiment of FIG. 15 additionally illustrates an intermodgenerator IMGTx_(m) (1510), where m may be from 1 to M, which may beused to estimate IM signal components caused by the Tx PA in chain m. Itis expected that the weights used for any CFIR3 following the IMGTx_(m)would be the same as for the other CFIR3 under the reasonable theorythat the IM components out of the Tx_(m) would experience the samepropagation paths to each Rx_(n) as the rest of X_(m)(t). Each IMGTx_(m)(1510) may also take from the ZCLCG (675) certain weights (not shown)that optimize parameters within the intermod model based on observed orcalculated results.

In addition to IMGTx_(m), other intermods of potential interest includethose created in the RF LNA of each receive chain n and those createdwithin the other cancellers (such as L1 per the indication above). EachRF LNA may generate intermods of all M Tx signals each of which may beuniquely transferred to each Rx_(n). Thus, cancellation of suchintermods requires at least a bank of M CFIR3Rx_(m) (1502-m) for eachTx_(m), each comprising N CFIRs of length D3RF taps and weights (notshown). The intermods are then modeled based on M inputs at eachrespective IMGRx_(n) (1504-n) wherein certain weights (not shown) may bepassed to optimize the model in view of observed results. Similarly, forintermods created within each canceller branch of L1 (or of L2, notshown), these can be estimated for each X_(m)(t) by applying Tx_(m)(t)to IMGC1_(m) (1512) as shown above and then applying the result to abank of N CFIR2C1-m each of D3C1 taps in length with weights (not shown)supplied by the ZCLCG (675).

In specific embodiments, the various operational and design parametersare chosen such that none of the intermod cancellers depicted above arenecessary. However, this will not always be the case particularly forhigh transmit powers associated with longer range operation.

Note the preceding discussion shows both C3R_(n) and C3D_(n) but inpractice only one of these would normally be present from a single CFIR3bank as depicted. If it were desired to have both, then there would needto be additional CFIR circuitry present (not shown) to account for thedifference in magnitude, phase and delay between the two types ofoutputs in order to effect cancellations, or alternatively the CFIR3bank would be replicated separately (i.e. CFIR3R & CFIR3D) with separateweights (not shown) for each to effect a “L3R” and a “L3D” in parallel.

All of the CFIR, IMG and summation circuits (1514, 1520, etc) depictedin FIG. 15 can be realized by conventional digital circuit techniques inan ASIC or FPGA, or alternatively by software in a DSP. The optional C3Rupconverter chains (1506-1 to 1506-N) depicted can be implemented in anRFIC or in commercially available RF transmitter components.

FIG. 16 is a block diagram of a portion of a ZDD enabled IBR including aZDD Canceller Loop Coefficients Generator (ZCLCG) according to oneembodiment of the invention. Note that the architecture depicted in FIG.16 may be implemented in many different ways as hardware, software on aprocessor, or a combination of both. In specific embodiments, the ZCLCGcan be realized within commercial of the shelf processors, or in ASIC orFPGA (in whole or in part).

In one embodiment, a basic process of subsequent cancellations based onan RRC setting (first order) and RLC setting (second order) is employed.In one exemplary embodiment of a ZDD enabled IBR, a start-up techniqueis for both ends of an IBR link in a given channel to start-up initiallyin a TDD service mode that allows the IBRs to exchange key RRC data andestablish common frame sync.

Then, in one embodiment based on RRC, L0 “coefficients” (possiblyembodied as just selectable settings) are chosen first, typically from anon-volatile memory set by design or factory calibration (and possibly,or alternatively, within the Coefficients Memory (1606)). To the extentthat such L0 coefficients or settings have multiple valid values forgiven RRC parameters, such values can be tested by inserting Tx-m chainpreambles (serially or in parallel as described below for L1), and thenapplying FFT (or even just an RSSI, not shown, but available from thesampling receiver chain (1225-n) in one embodiment, or elsewhere infurther embodiments) to each (or all) Rx_(n) to measure the effects onundesired Tx_(m) signal leakage into each Rx_(n). This can be iterateduntil a minimum leakage value is determined. Alternatively, the L0settings may be balanced with an identified desirable IBR receivesignal. In one embodiment, such a balance comprises determining settingsachieving a target threshold receive signal strength, or SNR, or othermetric for a desired receive signal, and a target L0 isolation level. Insuch an embodiment, compromises between receive desired SNR (or othermetric), and L0 isolation requires balance and may be achieved byoptimizing the value (maximum or minimum) of a formula defining a metricsuch as the following:

Vo=max(f _(DS)(W ₀(i))*f _(ISO)(W ₀(i))), for all i,

In the equation above, f_(DS)(W₀(i)) is a function of the desiredreceive signal, where W₀(i) are L₀ coefficients sets as a function of i,and in one embodiment include the RRC antenna selection settings of theIBR antenna array (648). Note that i is an index which ranges in valuefrom 1 to the total number of possible W₀(i) setting combinations.

In the equation above, f_(ISO)(W₀(i)) is a function of the isolationbetween each transmit chain and each receive chain, where W₀(i) are L₀coefficients sets as a function of i, and in one embodiment include theRRC antenna selection settings of the IBR antenna array (648).

The functions f_(DS), and f_(ISO) in some embodiment are linearfunctions, while in other embodiment are nonlinear functions, or acombination. In one embodiment, the functions are a comparison to one ormore thresholds, or fuzzy logic processing. In other embodiments thefunctions take the form of a global optimization function determiningthe maximum link throughput between two or more IBRs. Embodiments ofsuch a function balance the target IBR received throughput andreliability and the current IBR's receive throughput and reliability asfunction of W₀(i) settings. Examples of some of such embodiedoptimization algorithms associated with the L0 setting process, orassociated with the other loop settings may be found in SystemsEngineering in Wireless Communications, by Koivo and Elmusrati (ISBN0470021780).

Second, in one embodiment holding the L0 coefficients constant, basedalso on RRC primarily, an initial guess for the L1 coefficients would beread from a non-volatile memory set by design or factory calibration(and possibly within the Coefficients Memory (1606)). An iterativeprocess would commence wherein Tx_(m) preambles (either serially or inparallel initially) are inserted into each Tx_(m) and the effect on eachIQ magnitude bin for each Rx_(n) is considered to modify thecoefficients to minimize observed Tx_(m) in each Rx_(n). There arenumerous techniques that the Coefficients Generator Processor (1610)and/or Coefficients Calculator (1608) can use, such as iteration bysteepest descents (or alternatives such as those disclosed within Koivoand Elmusrati), to determine the coefficients in view of the previousvalues of W1_(m,n,d) and previous IQ magnitude bins given a new observedset of IQ magnitude bins for a new tested set of W1_(m,n,d). Suchprocessing may be performed as time domain based algorithm, or utilizingfrequency domain based processing, or as a combination of both domains.In some embodiments, the Tx_(m) preambles may be inserted in parallel tooptimize the W1_(m,n,d) in view of Tx_(m1) to Tx_(m2) leakage thatpasses through L1 and creates an additional leakage path independent ofL0 into each Rx_(n) from each Tx_(m). In an exemplary embodiment, theCritical Timing Unit (1616) schedules the preamble insertions and FFTsampling in view of the Tx Symbol Clock (for precise preamble symbolinsertion) and the Tx Frame Sync (so that, for example, Tx_(m) preamblesare sent when other IBRs in receive antenna view of the instant IBR inthe same Rx channel have their Tx signals substantially inhibited inpower). Such timing may also be coordinated via the IBMS Agent (700) incooperation with other IBRs or an IBMS Server as described in co-pendingapplication U.S. Ser. No. 13/271,057 by a common inventor and assignee.

In an embodiment where L1 is not present but L2 is, the above processwould be performed in a similar manner for L2 in some embodiments. Tothe extent that the delay range of such an L2 includes long delaycancellations, the optimization of such W2_(m,n,d) would in someembodiments, be performed similar to the process described below for L3in some embodiments. Alternatively or in addition, the L2 coefficientsmay be calculated by closed form approaches rather than using iterativealgorithms. Such algorithms may include so called MMSE and LMS basedapproaches in some embodiments, which will be described in more detail.

After the L1 coefficients have converged, reflection cancellations aremade using either L2 or L3 (for example, using L3, as in L3D only incertain embodiments as depicted associated with FIG. 17, or parallel L3Rand L3D loops in other embodiments). In some embodiments the calculationof L3 coefficients would also be an iterative process to determineW3_(m,n,d) coefficients but it is unlikely that a meaningful initialguess would exist from design or factory calibration relative to anactual deployment location. Thus, the number of preamble/FFT cycles toconverge may be substantial. Also, the number of calculations requiredfor each preamble/FFT cycle in view of the many potential CFIR taps inL2 or L3 may be substantial. In the case where desired Tx power andcertain component parameters causes substantial IM signals that areinsufficiently cancelled by L1 and/or L2, then a final process tocompute CFIR3 tap weights and IM model parameters is required. In oneembodiment, this could be done by having the Critical Timing Unit (1616)also control a Tx Power Control circuit (1602) (for example, shiftingaway 1 or 2 bits in each Tx chain for symbol timing level 6 or 12 dBpower control steps) that should allow FFT sampling to pick up theeffects of IM processes due to the non-linear effect of such leakagesignals within each Rx-n as a function of Tx power level.

The above process assumes CC-ZDD wherein some or all of the Tx-m channeloverlaps with the Rx-n channel. For CB-ZDD wherein the Tx and Rxchannels are adjacent, it may be preferable to use the above approachwith the Rx channel BW in the downconverter chains temporarily set tocover some or all of the Tx and Rx together to more efficientlydetermine various cancellation coefficients. For CB-ZDD with disparateTx and Rx channels, it may be preferable to temporarily force the RRC totune the signal receiver chains (1225-n) to the Tx channel so thatvarious cancellation coefficients can be efficiently determined from theTx leakage signals. However in CB-ZDD, even if neither of the aboveoptions is exercised, it should still be possible to determinecancellation coefficients simply from Tx noise (and out of channel IMsignals if present) using substantially the same procedure describedabove for CC-ZDD.

After both ends of a particular link have converged to an initiallyacceptable set of canceller coefficients, it is also necessary to updateand maintain these coefficients in view of changing environmentalparameters such as temperature or voltage internal to each IBR orchannel obstructions external to each IBR. It is expected that L0parameters would not be part of this update process. One exemplaryprocedure for this update process would be to periodically, such as onceevery L3_(U) frames (where L3_(U) may be 1 to 10 depending on theenvironment), have the Critical Timing Unit send one or more Tx_(m)chain preambles, while the other IBRs in view of the Rx antennas at theRx channel frequency substantially inhibit their Tx power, andeffectively repeat the W3_(m,n,d) coefficient calculation process if theIQ magnitude bins have increased beyond some threshold from the previoustest stored in memory. Similarly, every L1_(U) frames (where L1_(U) maybe 100-1000 depending on the environment), the W1_(m,n,d) coefficientsmay be re-tested with C2 and/or C3 inhibited to recalculate W1 if achange beyond a threshold has occurred. If so, recalculation of W2and/or W3 (and/or IM parameters) would need to follow. Alternatively,rather than every L1_(U) frames, this W1 test may be scheduled only ifthe L3 (and/or L2) update process fails to keep the residual Tx signalsin one or more Rx_(n) below a particular threshold value. In some ZDDIBR installations, the channel dynamics may be far slower or fasterchanging than in others. Thus, it may also make sense to have L1_(U),L2_(U), and/or L3_(U) into parameters that are adaptive to the estimatedchannel dynamics. For example only, if current values of L1_(U), L2_(U),and/or L3_(U) consistently produce insubstantial changes to theirrespective W1, W2, and/or W3 coefficients, then increase the givenvalues of L1_(U), L2_(U), and/or L3_(U) until some predetermined maximumis reached. Conversely, if substantial changes to coefficients doresult, then reduce the given values of L1_(U), L2_(U), and/or L3_(U)until some predetermined minimum is reached.

As will be discussed, in other embodiments a closed form LMS or MMSEcalculation may be made to determine L2 or L3 weights (or other weightsfor that matter).

In one embodiment associated with calculating L2 weights, a leastsquares closed form based approach may be utilized in a C2D (oranalogously for calculating L3 weights in a C3D) cancellation whereinthe sampled RF Transmit Reference Signals (X_(S1) to X_(SM))corresponding to signals associated with 680-1 through 680-M, usingsampling receivers (1406-m), are compared with the receive chain outputsignals (674) prior to summer (or alternatively referred to ascombiners) 1212-n (or at 1212-n with C2D and/or C3D temporarilyinhibited) to calculate the coefficients W2_(m,n,d), utilizing a closedform calculation. Some embodiments will utilize a least squarescalculation, or MMSE calculation which when performed in the time domainrequires a large matrix inversion. Alternatively such calculations maybe performed in the frequency domain to achieve a closed form solution.For example in one embodiment, the following process is followed foreach receiver chain n, where n varies from 1 through N:

1) A vector of sampled receive data (Zs_(n)) is assembled from S_(R)complex valued time domain samples of Z_(Sn)(t) at the output ofreceiver chain 1225-n, taken from the input (674-n) of cancellationcombiner (alternatively referred to as cancellation summers) 1212-n.Note that Z_(Sn r) is arranged to have the dimensions S_(R) by 1, andthat S_(R)≧D2, where D2 is the number of taps in each CFIR2 (1408).Next, the Fourier transform of Zs_(n) is taken to provide FZs_(n) (alsohaving dimensions S_(R) by 1). Such a Fourier transform may be realizedin any one of known conventional approaches, including utilizing FFTprocessing block 1614, with additional coupling to signals at referencepoints 674 to obtain time domain samples of Zs_(n) (not shown in FIG.16) or from Rx_(n) directly with C2D and/or C3D temporarily inhibited.

2) M vectors are respectively assembled for each block 1404-m coupled toRF Transmit Reference Signals 680-m, where m=1 through M. Each vector(Xs) of sampled RF Transmit Reference Signals data is assembled fromS_(R) complex valued time domain samples of X_(m)(t) at the output(1409-m) of the sampling receiver chain (1406-m) of block 1404-m, foreach reference signal. Note that each vector Xs_(m) is arranged to havethe dimensions S_(R) by 1, and that S_(R)≧D2, where D2 is the number oftaps in each CFIR2 (1408). Next, the Fourier transform of Xs_(m) istaken to provide FXs_(m) (also having dimensions SR by 1). Such aFourier transform may be realized in any one of known conventionalapproaches, including utilizing FFT processing block 1614, withadditional coupling the signals at reference points 1409-m of eachsampling receivers 1406-1 through 1406-M to obtain time domain samplesof Xs_(m) (not shown in FIG. 16). The M vectors of FXs_(m), where m=1through M, may further be arranged to form a matrix FXs of dimensionS_(R) by M, where each column contains a single vector FXs_(m).

3) Next, in the frequency domain, and on a bin-by-bin basis, a LeastSquares estimation is performed. For a reference on the least squaresestimator see eq. 3.33, Smart Antennas For Wireless Communications,Rappaport; ISBN 0-13-71987-8 and derivation 3 L, Linear Algebra and itsApplications, Strang; ISBN 0-15-551005-3. Also see “Least SquaresProblems with several Variables” as one example utilizing complexmathematics for the current embodiment, though alternative solutions arecontemplated. Adapting the MMSE and Least Squares approach to thecurrent application yields Eq. 14-1.

FW2_(n) ^(EST)(b)=[FXs(b)^(H) FXs(b)]⁻¹ FXs(b)^(H) FZs _(n)(b)  (Eq.14-1)

where b indexes frequency bins 1 through S_(R), and where the “H”superscript refers to the Hermitian (conjugate) transpose of a matrix,and the superscript “−1” refers to a matrix inversion and, where,FZs_(n)(b) is a complex scalar and FXs(b) is comprised of a 1 by Mvector, and FW2_(n) ^(EST) (b) is an M by 1 vector, where each elementcomprises the b^(th) complex frequency domain bin value of the frequencydomain estimation of the filter taps associated with CFIR2_(m,n).

4) Following computation for all frequency bins from b=1 through S_(R),the newly determined (M by S_(R)) matrix FW2^(E) _(n) ^(EST) may becombined utilizing a weighted averaging, or other filtering approaches,on an bin by bin basis with previously utilized or estimated versions ofmatrix FW2_(n) ^(EsT) resulting in matrix FW2_(n) ^(EST) Alternatively,the frequency domain based filtering may be performed in a later timedomain step, and vector FW2_(n) ^(EST) may be substituted for thesubsequent step referencing the matrix FW2_(n) ^(Filt).

5) Next an inverse FFT of FW2_(n) ^(Fult) is performed on a row wisebasis (over SR samples, for each row m) resulting in W2_(n) ^(EST) whichcomprises estimates of the time domain FIR filter weights as a vector ofdimension M by S_(R), where S_(R) is ≧D2 (the number of filter taps perFIR2 (1408)).

6) Finally W2_(n) is calculated by performing a truncation of W2_(m)^(EST) from M by SR to a M by D2, where for each receive chain the mostsignificant magnitude coefficients will be contained in the first D2values of the M rows of W2_(n) ^(EST). Additionally, a time domainsample-by-sample averaging, weighted averaging (FIR), recursive (IIR)filtering may be performed with previous calculated versions of W2_(n)^(EST) or W2_(n). Note that each row W2_(n) corresponds to the weights,W2_(m,n,1) to W2_(m,n,D2) for the corresponding CFIR_(m,n) (1408). Notethat this processing equally applies to the calculation of the L3coefficients as well, with the exception that such calculation beprocessed on the output of the cancellation of C2 (if present), and onlywhile the W1, W2, and W3R weights (if present) are held static.Additionally the transmit reference signals would be based upon the puredigital reference signals Tx1 . . . M (672) input to the transmitchains, rather the RF Transmit Reference Signals (680).

Referring now to FIG. 17, a block diagram of a portion of a ZDD enabledIBR including Loop 1 (C1) (1204) and Loop 3 (C3D) (1208) cancellers isdepicted according to one embodiment of the invention. Note that alllabeled signals in FIG. 17 are complex, represented by either I and Q ormagnitude and phase. In one embodiment, the isolation in L0 (1202) andthe cancellation capability of L1 are sufficient that only L3 withdigital baseband outputs is required for CC-ZDD operation. Furthermorewith careful design, the L3 (1208) in a preferred embodiment does notneed IM cancellation circuits.

Note that L0 (1202) includes the effects of each transmit antenna signalX_(m)(t) on every receive antenna signal Y_(q)(t), as well as on everyother X_(m)(t). Note also that every antenna, X_(m)(t) and Y_(q)(t), maybe subject to receiving unwanted interference I(t) as well as thedesired signal from another IBR and various undesired leakage signalsfrom the instant IBR.

The goal of L0, L1 and L3 is that each Rx(t) does not have a level ofany X_(m)(t) (or other undesired signal introduced by the instant IBRand not the outside world radiating into each receive antenna) thatcauses appreciable (typically more than 1 dB) desensitization at thedemodulator relative to an equivalent receiver where each X_(m)(t) atthe instant IBR has zero power.

Referring now to FIG. 18, a block diagram depicting of a portion of aZDD enabled IBR including a Loop 2 (C2D) and Loop 3 (C3R) cancellersaccording to one embodiment of the invention. In embodiments associatedwith FIG. 18, wherein C2D of FIG. 14, and C3R of FIG. 15 are utilized,the noise from the cancellation transmitters 1506-1 to 1506-N may becompensated for by sharing L0 frequency sources (VCOs) between thesignal transmitters and the C3R transmitters such that the phase noiseis common to all up converted signals allowing for the cancellation ofthe transmitted phase noise at summers (alternatively referred to ascancellation combiners) 1210-n. Additionally, utilizing a shared VCObetween the C2D sampling receivers (1406-m) and signal receive chains(1225-n) allows for a common phase noise component between all receiversallowing for the cancellation of phase noise at baseband in C2D orrelated digital cancellation. Performing additional cancellation of thedigital base band transmit signals (Tx₁(t) and Tx₂(0), the individualphase noise components may be recovered for use in further cancellationprocessing using noise components individually (the shared transmitphase noise, and the shared receive phase noise). As a result,transmitter leakage signals from IBR transmit array (648A) may becanceled at 1210-n utilizing C3R processing allowing for a cancellationof the leakage signals such that the ADCs within the signal receivechains (1225-n) are not saturated. Such embodiments sharing a common VCOamong all up converters and a common VCO among all down converters downconverters allows for the cancellation of the transmitter leakage signaland the transmitter phase noise at 1210-n. In other embodiments,transmitter noise may not be cancelled when the transmitter leakagesignal is cancelled (when the C3R up converter VCOs are not shared withthe transmit chain VCOs) and remain in the signal receive chains(1225-n), and the remaining phase noise may be cancelled in C2Dprocessing and associated processing utilizing references for the upconverter or down converter phase noises.

However, in some embodiments utilizing commercially available chip setimplementations, VCOs are shared only among pairs of transmitters andpairs of receivers. As discussed previously, the phase noisecombinations may be recovered and compensated for in furthercancellation processing at base band during C2D or C3R processing.

Referring now to FIG. 19 through FIG. 24 a detailed mathematicaldescription of one the embodiments of FIG. 17 utilizing L1, and L3 arepresented. Note that within the references figures all multiplies “x”shown are complex, though realizations of such complex multiplies may beperformed by analog structures such as mixers or by digital structuressuch as combinatorial logic gates (i.e. “XOR”) or adders.

FIG. 19 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR transmitter according to one embodiment ofthe invention. Referring to (19-1) and (19-2) note that X_(m)(t) islinearly related to Tx_(m)(t) except for the additive IM and noiseterms. The IM terms may be modeled from Tx_(m)(t) depending on themechanism that causes them. The noise is uncorrelated from Tx_(m)(t) andthus L3 cannot cancel the Tx noise in the receiver in embodiments ofFIG. 17. Unless L1 (or theoretically L2, but subject to practicallimitations described earlier) cancels this noise to a level acceptablybelow each receive chain's internal noise level (plus receiveduncorrelated noise/interference), then this Tx noise will desense thedemodulator and limit the overall link margin in this embodiment.

The sampled Tx_(m)(t) signals utilized by L1 are predominantlyG_(Sm)×X_(m)(t). However, they may also include unwanted leakage ofother transmit signals on a theoretically infinite number of propagationpaths as shown in Eqs. 19-3 and 19-4 of FIG. 19. There may also beuncoordinated interference present at each Tx antenna but generally ifthis were significant it would probably already cause significantinterference at one or more receive antennas in specific embodiments.There is also at least thermal noise present at the input to L1.Although this noise N_(Sm)(t) may be far below the transmitter noise[G_(Sm)×N_(Txm)(t)] in magnitude, the effect of N_(Sm)(t) combined withthe transfer function within L1 and its noise figure will be injected atZ_(n)(t) into each receiver chain, while the transmitter noise istheoretically cancelled out (and there is no practical way to cancelN_(Sm)(t)).

FIG. 20 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR receiver according to one embodiment of theinvention. At the receiver each Y_(q)(t), of Eq. 20-1 receives a desiredsignal from its peer IBR (Y_(Rxq)(t)) as well as both exemplary transmitsignals X_(m)(t), each of which reaches Y_(q)(t) by a theoreticallyinfinite number of delay paths and complex transfer functions, plus theinput noise and interference at each antenna.

At the inputs to each downconverter chain, Z_(n)(t) (Eq. 20-2), thesignal is composed of all 8 Y_(q)(t) signals multiplied by respectivecomplex transfer functions and subject to a small delay plus theequivalent input noise for the downconverter chain and the L1cancellation signal, as well as any intermod products generated in thecircuits from antenna q to downconverter n (especially in view of Txleakage from X_(m) that may be much larger than Y_(Rxq)). Note thatwithin 648B the effect of the switch matrix and the LNA power down whena particular receive antenna is not selected typically results in theselected q_(n) mapping having a magnitude of G_(Rxq,n) that greatlyexceeds that of all unselected mappings.

The L1 cancellation signal can be described as a summation across thetwo sampled transmit signals Xs_(m) (which each comprise X₁ and X₂ dueto finite Tx to Tx antenna isolation) weighted by complex transferfunctions W1_(m,n,d) at delays of approximately “d*Δt₁” (but notnecessarily uniform delay steps in some embodiments) and G1_(m,d) (insome embodiments these might all be unity but in practice with RF delaylines these will vary somewhat in other embodiments), plus undesiredintermodulation products created within L1 as indicated.

FIG. 21 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR receive chain input according to oneembodiment of the invention.

This representation of Z_(n)(t) (Eq. 21-00) illustrates five constituentcomponents for each Rx downconverter input chain signal:

(21-01) This is the “desired Rx chain signal” component that will beeventually channel multiplexed, equalized, and demodulated in thebaseband Rx PHY. This is dominated by the switch matrix selection from qto n.

(21-02) This is the “Rx chain input noise floor” (which includesinterference at the Rx antennas) component that leads to finite SINR forthe receive chain. This should also be dominated by the switch matrixselection from q to n in consideration of the thermal noise floor and RFfront-end noise figure. Ideally the dominant G_(Rxq,n)×N_(Rxq) is muchgreater than N_(DCn) as well in good RF design practice.

(21-03) This is the “undesired Tx leakage” component that reflects thecumulative effects of finite isolation in L0 from any Tx antenna m toany receive antenna q in view of the switch matrix selections.

(21-04) This is the L1 cancellation component (or “L1 output signal”)which in an ideal world would be exactly equal to, but opposite in signto, the undesired Tx leakage component. In reality, this is notpossible, but if L1 sufficiently cancels certain key sub-components ofthe Tx leakage, then L2 or L3 may be able to sufficiently cancel therest for CC-ZDD. In the case of CB-ZDD, then L1 needs to cancel only theTx leakage noise (and IM signals if present) in the Rx channel to beacceptably low relative to the noise plus interference component.

(21-05) This is the “undesired Rx intermods” (of the Tx leakage signals)produced in the RF front-end (typically the LNA) of each path from Rxantenna q to downconverter input n. Ideally, the L0 isolation and thelinearity of the LNA and/or switch matrix would cause such intermods tobe acceptably low relative to the noise plus interference component bydesign such that this component is negligible. It is generally notpossible to cancel this particular intermod component by L1. However, insome cases it may be feasible to model the generation of this intermodcomponent relative to the Tx_(m) digital baseband signals and thuscancel it in L3.

FIG. 22 is a diagram of a further mathematical representation ofdepicted signals at a ZDD enabled IBR receive chain input according toone embodiment of the invention. Substituting previous representationsof X_(m)(t) (Eq. 19-1 and 19-2) and C1_(n)(t) (Eq. 20-3) creates a moredetailed description of Z_(n)(t) (Eq. 22-00).

This representation of Z_(n)(t) now illustrates nine constituentcomponents for each Rx downconverter input chain signal:

Equation components (22-01) and (22-02) are equivalent equation segments(21-01) and (21-02) of FIG. 21, respectively.

Equation Segment (22-03) is the first of three subcomponents of the“undesired Tx leakage” component from the previous slide. Thisparticular subcomponent is representable in terms linearly related tothe original Tx chain baseband digital signal Tx_(m)(t).

Equation Segment (22-04) is the second of three subcomponents of the“undesired Tx leakage” component from the previous slide. Thisparticular subcomponent comprises the Tx chain output noise.

Equation Segment (22-05) is the third of three subcomponents of the“undesired Tx leakage” component from the previous slide. Thisparticular subcomponent comprises the Tx chain output intermods.

Equation Segment (22-06) is the first of three subcomponents of the “L1output signal” component from the previous slide. This particularsubcomponent is the actual L1 cancellation signal that is complicated bythe finite Tx-Tx antenna isolation as represented in L0tx_(m1,m2,i).This “L1 cancellation signal” would ideally cancel all three of theundesired Tx subcomponents.

Equation Segment (22-07) is the second of three subcomponents of the “L1output signal” component from the previous slide. This particularsubcomponent comprises the interference at the Tx antennas and the noisegenerated by L1. It is not possible to cancel the noise term in thissub-component. Therefore, L1 and the Rx front-end must be designed suchthat the noise term in this sub-component is acceptable relative to the“Rx chain input noise floor” component. In practice, this involvesminimizing the noise output from L1 and raising the gain in the Rxfront-end without causing other problems (such as downconvertersaturation or Rx front-end undesired intermods). It is also not possibleto cancel the interference term in this subcomponent at least fromwithin the ZDD subsystem of the IBR. However, this interference I_(Txm)is likely correlated to I_(Rxq) in practice and thus an IBR withredundant Rx chains may be able to cancel such interference in the Rxchannel multiplexer. Note that in practice, W1×G1×Gs_(m) is likelysubstantially lower in magnitude that G_(Rxq,n) for the selected q of agiven n.

Equation Segment (22-08) is the third of three subcomponents of the “L1output signal” component from the previous slide. This particularsubcomponent arises from intermods of X_(m)(t) (or effectively Tx_(m)(t)if the EVM is small) generated within L1. Preferably, these intermodsare acceptably low relative to the “Rx chain input noise floor”component by design and hence effectively negligible. In some cases, itmay be feasible to model the generation of this intermod componentrelative to the Tx_(m) digital baseband signals and thus cancel it inL3.

Equation Segment (22-09) is equivalent to Equation Segment (21-05) ofFIG. 21.

FIG. 23 is a diagram of a further detailed mathematical representationof depicted signals at a ZDD enabled IBR receive chain input accordingto one embodiment of the invention. Further substituting previousrepresentations of X_(m)(t) into the L1 cancellation signal creates aneven more detailed description of Z_(n)(t) (Eq. 23-00 of FIG. 23).

This representation of Z_(n)(t) now illustrates eight constituentcomponents for each Rx downconverter input chain signal:

Equation components (23-01) and (23-02) are equivalent equation segments(21-01) and (21-02) of FIG. 21, respectively.

Equation component (23-03) is the first of three subcomponents of the“residual Tx leakage” component which is expressed in view of the threesub-components of the “undesired Tx leakage” in combination withanalogous sub-components of the L1 cancellation signal of the previousslide. This particular subcomponent is representable in terms linearlyrelated to the original Tx chain baseband digital signal Tx_(m)(t)(across all Tx chains). To the extent that the magnitude of[G_(Txm)×Tx_(m)] times L0_(m,q,i) exceeds the equivalent input noise(plus interference) for any [t_(m,q,i)+t_(Rxq,n)]>[t_(sm)+t_(D1)], thenat least some L3 cancellation will be required or the receive chain willbe desensed.

Equation component (23-04) is the second of three subcomponents of the“residual Tx leakage” component. This particular subcomponent comprisesthe residual Tx chain output noise (with contributions from all Txchains). It is imperative (in some embodiments) that t_(D1) be chosensuch that [t_(sm)+t_(D1)]>[t_(m,q,i)+t_(Rxq,n)] for any “i” wherein themagnitude of [N_(Txm)×L0_(m,q,i)] exceeds the equivalent input noise(plus interference) of each receive chain and further that D1 be largeenough in number of taps such that any frequency dependence of theeffective transfer function of L0 can be sufficiently approximated. Thisis necessary because L3 in this exemplary embodiment cannot cancel anyresidual Tx noise in a receive chain.

Equation component (23-05) is the third of three subcomponents of the“residual Tx leakage” component. This particular subcomponent comprisesthe Tx chain output intermods (with contributions from all Tx chains).Ideally in the current embodiment, t_(D1) would be chosen such that[t_(sm)+t_(D1)]>[t_(m,q,i)+t_(Rxq,n)] for any “i” wherein the magnitudeof [X_(IMT) _(xm) ×L0_(m,q,i)] exceeds the equivalent input noise (plusinterference) of each receive chain and D1 would be large enough innumber of taps such that any frequency dependence of the effectivetransfer function of L0 can be sufficiently approximated so that furthercancellation of Tx IM products in L3 of this exemplary embodiment is notrequired.

Equation components (23-06, 07, and 08) are equivalent to (22-07, 8, and9) of FIG. 22.

Note that the relationship of W1_(m,n,d) to G_(Rxq,n), G1_(m,d), G_(Sm),L0tx_(m1,m2,1), L0_(m,q,i), and the delays is theoretically identicalfor Tx_(m)(t), N_(Txm)(t) and X_(IMTxm)(t). Thus in CC-ZDD solving forW1_(m,n,d) such that the residual Tx signal is minimized should alsominimize the residual Tx noise and intermods at a given Rx chain inputn. For CB-ZDD, this is still true for residuals in the Tx channel, butnot necessarily true for residuals in the Rx channel if W1_(m,n,d) wasdetermined by minimizing residual Tx signal in the Tx channel. Inparticular, for non-overlapping CB-ZDD, the frequency dependence of L0may make the W1_(m,n,d) that minimizes the residual Tx noise in the Rxchannel somewhat different than that which minimizes residual Tx noise(or residual Tx signal) in the Tx channel. However, determiningW1_(m,n,d) may be far simpler in the Tx channel by minimizing residualTx signal so this may be a good interim step despite the likelyrequirement of temporarily operating the Rx chain at the Tx channel (andnote that to the extent iteration is used, the first guess may stillpreferably come from a table set by factory calibration and/or design asdiscussed previously). Final optimization of W1_(m,n,d) in CB-ZDD mayrequire processing gain by averaging or multiple iterations across manyresidual signal tests due to the statistical variations of the noise.

FIG. 24 is a diagram of the mathematical representation of depictedsignals at a ZDD enabled IBR receive chain digital output signalsaccording to one embodiment of the invention. Consider now the Rx chainoutput in the digital domain (24-01) which is linearly related to inputplus the L3 cancellation signal. For the specific case where all RFfront-end and L1 intermods and noise are sufficiently small relative tothe receive chain noise floor by design, and the residual Tx noise andTx intermods are sufficiently cancelled by L1 relative to the receivechain noise floor, then the receive chain digital output signals can bewritten approximately as composed of three components described asfollows:

The first component is the desired Rx chain signal (24-11). The secondcomponent (24-12) is the Rx chain noise floor plus interference fromsources other than the instant ZDD IBR. Because L1 connects the Txantennas in a path to each receive chain independent of the Rx antennas,the interference at the Tx antennas is also in each receive chain perthis component. However, if I_(txm) is correlated with I_(Rxq), then incertain receive architectures such interference may be cancellable bythe Rx FDE or other mechanism within the Rx channel multiplexer assumingredundant Rx chains.

The third component (24-13) is the residual Tx signal (withcontributions from all Tx chains due to Tx-Tx antenna coupling). The L3cancellation signal must be optimized by finding W3_(m,n,d3)coefficients such that this residual Tx signal is sufficiently smallrelative to the Rx chain noise floor in CC-ZDD. Because the residual Txsignal is linear with respect to Tx_(m)(t), then for sufficiently largeD3 and t_(D3)>t_(m,q,i) for any L0_(m,q,i) of interest, it should alwaysbe possible to find W3_(m,n,d3) such that this residual Tx signal issufficiently minimized. For non-overlapping CB-ZDD, this component maynot be important as the frequency dependence of G_(DCn) may greatlyattenuate this component relative to the desired Rx signal and the Rxnoise floor, and the residual may be effectively discarded by the FFT inthe Rx channel multiplexer.

One or more of the methodologies or functions described herein may beembodied in a computer-readable medium on which is stored one or moresets of instructions (e.g., software). The software may reside,completely or at least partially, within memory and/or within aprocessor during execution thereof. The software may further betransmitted or received over a network.

The term “computer-readable medium” should be taken to include a singlemedium or multiple media that store the one or more sets ofinstructions. The term “computer-readable medium” shall also be taken toinclude any medium that is capable of storing, encoding or carrying aset of instructions for execution by a machine and that cause a machineto perform any one or more of the methodologies of the presentinvention. The term “computer-readable medium” shall accordingly betaken to include, but not be limited to, solid-state memories, andoptical and magnetic media.

Embodiments of the invention have been described through functionalmodules at times, which are defined by executable instructions recordedon computer readable media which cause a computer, microprocessors orchipsets to perform method steps when executed. The modules have beensegregated by function for the sake of clarity. However, it should beunderstood that the modules need not correspond to discreet blocks ofcode and the described functions can be carried out by the execution ofvarious code portions stored on various media and executed at varioustimes.

It should be understood that processes and techniques described hereinare not inherently related to any particular apparatus and may beimplemented by any suitable combination of components. Further, varioustypes of general purpose devices may be used in accordance with theteachings described herein. It may also prove advantageous to constructspecialized apparatus to perform the method steps described herein. Theinvention has been described in relation to particular examples, whichare intended in all respects to be illustrative rather than restrictive.Those skilled in the art will appreciate that many differentcombinations of hardware, software, and firmware will be suitable forpracticing the present invention. Various aspects and/or components ofthe described embodiments may be used singly or in any combination. Itis intended that the specification and examples be considered asexemplary only, with a true scope and spirit of the invention beingindicated by the claims.

What is claimed is:
 1. A radio with adaptable radio frequency (RF)cancellation for reduction of one or more transmit signals at a receiverwithin the radio, said radio comprising: a plurality of transmit radiofrequency (RF) chains, wherein each transmit RF chain is configured toconvert from a respective one of a plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals; aplurality of adaptable RF transversal filter sets configured to convertone or more RF cancellation input signals derived from one or more ofthe plurality of transmit RF signals to a plurality of RF transmitcancellation signals, wherein each one of the plurality of adaptable RFtransversal filter sets comprises: one or more adaptable RF transversalfilters, wherein each adaptable RF transversal filter is configured tofilter a respective one of the one or more RF cancellation input signalsto provide a respective one of one or more adaptable RF transversalfiltered signals; and an RF filtered signal combiner configured tocombine the one or more adaptable RF transversal filtered signals withineach one of the plurality of adaptable RF transversal filter sets toproduce one of the plurality of RF transmit cancellation signals; aplurality of antenna elements, wherein each of a plurality of receive RFsignals is derived from signals received from at least one of theplurality of antenna elements, wherein each of the plurality of receiveRF signals comprises at least one or more transmitter related signals,and wherein each of the one or more transmitter related signals isderived from at least one of the plurality of transmit RF signals; aplurality of RF cancellation combiners, wherein each RF cancellationcombiner is configured to combine a respective one of at least one ofthe plurality of RF transmit cancellation signals with at least one ofthe plurality of RF receive signals to provide a respective one of aplurality of receive chain input signals; and a plurality of receiveradio frequency (RF) chains, wherein each receive RF chain is configuredto convert from a respective one of the plurality of receive chain inputsignals to a respective one of a plurality of receive chain outputsignals; wherein the radio is configured to adapt at least one of theone or more adaptable RF transversal filters to reduce a level of aninterfering signal component within at least one of the plurality ofreceive chain output signals, wherein the interfering signal componentis derived from at least one of the one or more transmitter relatedsignals, and wherein the radio utilizes a metric that accounts for atleast the interfering signal component.
 2. The radio of claim 1, furthercomprising: one or more demodulator cores, wherein each demodulator coredemodulates one or more receive symbol streams to produce a respectivereceive data interface stream; and a frequency selective receive pathchannel multiplexer to produce at least two receive symbol streams fromat least two of the plurality of receive chain output signals.
 3. Theradio of claim 1, further comprising: one or more modulator cores,wherein each modulator core modulates a respective transmit datainterface stream to produce one or more of a plurality of transmitsymbol streams; and a transmit path channel multiplexer to produce theplurality of transmit chain input signals provided to the plurality oftransmit RF chains from the plurality of transmit symbol streams.
 4. Theradio of claim 3, wherein the transmit path channel multiplexer is afrequency selective transmit path channel multiplexer.
 5. The radio ofclaim 3, wherein the transmit path channel multiplexer utilizes transmitbeam forming.
 6. The radio of claim 1, wherein the radio is configuredto insert transmit preambles that can be utilized at least to obtain anestimate of the level of the interfering signal component.
 7. The radioof claim 1, wherein the radio is configured to estimate of the level ofthe interfering signal component by using at least a Fast FourierTransform (FFT).
 8. The radio of claim 1, wherein the radio isconfigured to determine the metric from at least one of a least squaresor an MMSE calculation.
 9. The radio of claim 1, wherein the metriccomprises one or more of an RF transmit leakage metric, a transmitleakage metric, an RSSI metric, a metric involving a correlation with areference signal, a metric involving a correlation with a transmit chaininput signal, a metric derived from a receive RF chain, a metricassociated with a desired receive signal, a metric associated withisolation between certain receive antenna array elements and certaintransmit antenna array elements associated with the intelligent radio, ametric based upon measurements of signals, channel estimates, orcancellation performance, a receiver performance metric such as signalto noise ratio (SNR), bit error rate (BER), or frame error rate (FER),or a metric associated with the forward error correction (FEC) decoder.10. The radio of claim 1, further comprising: one or more selectable RFconnections, wherein each selectable RF connection is configured toselectively couple to at least one of certain of the plurality ofantenna elements to derive at least one of certain of the plurality ofreceive RF signals.
 11. The radio of claim 10, wherein at least one ofthe one or more selectable RF connections comprises circuitry to combinesignals received from at least two of the certain of the plurality ofantenna elements.
 12. The radio of claim 11, wherein the circuitry tocombine signals comprises at least an RF switch.
 13. The radio of claim1, wherein no channel bandwidth associated with any of the plurality oftransmit RF signals at a time of transmission overlaps with any channelbandwidth associated with any of the plurality of receive RF signals ata time of reception, wherein the time of transmission and the time ofreception can be coincident.
 14. The radio of claim 1, wherein at leastone channel bandwidth associated with any of the plurality of transmitRF signals at a time of transmission overlaps at least partially with atleast one channel bandwidth associated with any of the plurality ofreceive RF signals at a time of reception, wherein the time oftransmission and the time of reception can be coincident.
 15. A radiowith adaptable baseband cancellation for reduction of one or moretransmit signals at a receiver within the radio, said radio comprising:a plurality of transmit radio frequency (RF) chains, wherein eachtransmit RF chain is configured to convert from a respective one of aplurality of transmit chain input signals to a respective one of aplurality of transmit RF signals; a plurality of receive radio frequency(RF) chains, wherein each receive RF chain is configured to convert froma respective one of a plurality of receive RF signals to a respectiveone of a plurality of receive baseband sampled signals; a plurality ofantenna elements, wherein each of the plurality of receive RF signals isderived from signals received from at least one of the plurality ofantenna elements, wherein each of the plurality of receive RF signalscomprises at least one or more transmitter related signals, and whereineach of the one or more transmitter related signals is derived from atleast one of the plurality of transmit RF signals; a plurality ofreceive baseband cancellation combiners, wherein each receive basebandcancellation combiner is configured to combine a respective one of theplurality of baseband transmit cancellation signals with a respectiveone of the plurality of receive baseband sampled signals to provide arespective one of a plurality of receive chain output signals; and aplurality of adaptable baseband transversal filter sets, wherein eachrespective one of the adaptable baseband transversal filter sets isconfigured to receive one or more baseband cancellation input signalsderived from the plurality of transmit chain input signals and isconfigured to provide a respective one of a plurality of basebandtransmit cancellation signals to a respective one of the plurality ofreceive baseband cancellation combiners, wherein each adaptable basebandtransversal filter set comprises one or more adaptable basebandtransversal filters, wherein each of the one or more adaptable basebandtransversal filters is configured to filter at least one of the one ormore baseband cancellation input signals and to provide one of one ormore adaptable baseband transversal filtered signals; wherein eachadaptable baseband transversal filter set further comprises a basebandfiltered signal combiner configured to combine the one or more adaptablebaseband transversal filtered signals associated with each adaptablebaseband transverse filter set to provide one of the plurality ofbaseband transmit cancellation signals; and wherein the radio isconfigured to adapt at least one of the one or more adaptable basebandtransversal filters comprised within at least one of the plurality ofadaptable baseband transversal filter sets to reduce a level of aninterfering signal component within at least one of the plurality ofreceive chain output signals, wherein the interfering signal componentis derived from at least one of the one or more transmitter relatedsignals, and wherein the radio utilizes a metric that accounts for atleast the interfering signal component.
 16. The radio of claim 15,further comprising: one or more demodulator cores, wherein eachdemodulator core demodulates one or more receive symbol streams toproduce a respective receive data interface stream; and a frequencyselective receive path channel multiplexer to produce at least tworeceive symbol streams from at least two of the plurality of receivechain output signals.
 17. The radio of claim 15, wherein the radio isconfigured to insert transmit preambles that can be utilized at least toobtain an estimate of the level of the interfering signal component. 18.The radio of claim 15, wherein the radio is configured to estimate ofthe level of the interfering signal component by using at least a FastFourier Transform (FFT).
 19. The radio of claim 15, wherein the radio isconfigured to determine the metric from at least one of a least squaresor an MMSE calculation.
 20. The radio of claim 15, wherein the metriccomprises one or more of an RF transmit leakage metric, a transmitleakage metric, an RSSI metric, a metric involving a correlation with areference signal, a metric involving a correlation with a transmit chaininput signal, a metric derived from a receive RF chain, a metricassociated with a desired receive signal, a metric associated withisolation between certain receive antenna array elements and certaintransmit antenna array elements associated with the intelligent radio, ametric based upon measurements of signals, channel estimates, orcancellation performance, a receiver performance metric such as signalto noise ratio (SNR), bit error rate (BER), or frame error rate (FER),or a metric associated with the forward error correction (FEC) decoder.21. The radio of claim 15, wherein at least one channel bandwidthassociated with any of the plurality of transmit RF signals at a time oftransmission overlaps at least partially with at least one channelbandwidth associated with any of the plurality of receive RF signals ata time of reception, wherein the time of transmission and the time ofreception can be coincident.
 22. The radio of claim 1, wherein the radiois a backhaul radio.
 23. The radio of claim 15, wherein the radio is abackhaul radio.
 24. The radio of claim 1, wherein at least one of theplurality of antenna elements is a directive gain antenna element. 25.The radio of claim 15, wherein at least one of the plurality of antennaelements is a directive gain antenna element.
 26. The radio of claim 1,further comprising: a cancellation controller, wherein the cancellationcontroller is configured to adapt the at least one of the one or moreadaptable RF transversal filters and to utilize the metric that accountsfor at least the interfering signal component.
 27. The radio of claim15, further comprising: a cancellation controller, wherein thecancellation controller is configured to adapt the at least one of theone or more adaptable baseband transversal filters and to utilize themetric that accounts for at least the interfering signal component.